Co-authored by Bhargavi Nisarga, Systems Engineer at TI

In the first five installments of this series, we’ve helped you build a fully featured and network-connected smart thermostat. In this installment, we will focus on how you can best optimize power and energy consumption to extend the battery life of your thermostat.

A battery-powered wireless thermostat gives you “peel-and-stick” flexibility to use the product anywhere, even in the absence of pre-wiring for power or communication. However, if you don’t design your thermostat to have an extended battery life, users will have to change the batteries often, which will affect your product’s overall cost-effectiveness and convenience factor. Let’s take a look at how to address this challenge.

Here are some terms we’ll use in the context of power savings in this blog post:

• Current consumption refers to the current consumed by a module/device when operational, and is represented in amperes.
• Power consumption refers to the current consumed at a specific supply voltage level. Power = I × V and is represented in watts. This term is also interchangeably used with current consumption.
• Energy consumption depends on three factors: voltage, current and time. Energy = I × V × time. The total energy consumed includes the average current consumption based on the ratio of the time period during which the module/device is enabled or disabled, and the respective currents during those periods.

When you look at a microcontroller (MCU) data sheet, you might be startled – the current numbers may just seem too high for battery-powered applications. But here is how you need to read the numbers: the current numbers specified in a device data sheet show the current consumption assuming that the module is running with a 100% duty cycle. For an analog-to-digital converter (ADC), this means that you are sampling and converting data continuously, this is not realistic for most “real world” applications.

As we discussed in the second installment of this series, you will normally have longer intervals in between two samples; during this interval, the ADC will not be active. Because of its low duty cycle, the ADC will not consume much energy. So to calculate the average current consumption for all modules used in your application (like the central processing unit [CPU], ADC and comparator), you need to calculate the ratio between the on and off time periods and accordingly apply the current numbers specified in the data sheet. That’s how you will calculate the average current consumption. The maximum consumption at a given instant determines the peak current consumption. Normally, the battery will determine the peak-current limit for your application.

Most low-power MCUs offer different power modes to reduce the application’s overall power and energy consumption, typically supported by the following device power modes: active, sleep, deep sleep and shutdown mode. The power modes differ from each other depending on the clock sources available and the modules powered in that specific power mode. In general, lower power consumption in a specific power mode helps keep lesser functions functional and lowers the speed at which the modules operate.

Using MCU power modes to optimize current consumption

Let’s take a look at what techniques are used in different device power modes to enable power savings:

• Active mode has highest current consumption compared to other power modes, because the on-chip CPU is active. The faster the CPU frequency of operation, the higher the current consumption; therefore, a microampere/megahertz (uA/MHz) parameter represents this current number. On-chip peripherals typically support clock-gating techniques to disable transistor switching within the peripheral circuitry when it is not in use or not enabled, and in turn enable current savings.
• In sleep mode, the high-frequency on-chip clock oscillators remain running. Clock-gating only the CPU enables it to resume executing instructions on the next clock cycle following a wake-up trigger.
• In deep-sleep mode, the high-frequency on-chip clock oscillators are disabled; a low-frequency clock is enabled (if needed) to keep a few critical peripherals running, such as the real-time clock (RTC) or the interval timer. Upon a wake-up trigger, it takes a few microseconds to re-enable the oscillators before resuming active or sleep-mode operation.
• Deep-sleep and shutdown modes use power-gating techniques to power down most parts of the device and keep only a few peripherals (or groups of peripherals) powered to achieve sub-1µA-range device current numbers.

Keep in mind that the typical current numbers in a device data sheet are specified for operation at room temperature. Current consumption at higher temperatures (especially for sleep and deep-sleep modes) may be much higher compared to typical numbers due to higher leakage current at the transistor level. You will need to consider the maximum current numbers in the device data sheet when calculating current consumption over the product’s operating temperature range.

You should plan to optimize power consumption overall by grouping the on-chip peripherals that the device needs at different points of operation, and use the various power modes to power and clock only those modules that need to be active at a given instant. For example, when your smart thermostat is not actively measuring temperature, updating the liquid crystal display (LCD) or sampling the microphone input, the device can enter deep-sleep mode, where only the low-frequency clock is enabled for operations such as the RTC, periodic interval timekeeping (for temperature measurement) and capacitive touch sensing.

Figure 1: Figure 1a shows an application-level duty cycling profile for a smart thermostat where the temperature is sampled every t-temp-sample; samples are processed and data is transmitted over the network every t-temp-transmit. Increasing t-temp-sample and t-temp-transmit at the application level reduces the overall power consumption.

Figure 1b shows the corresponding device level power profile capturing device operation in various power modes including:

• Deep-sleep mode when waiting between temperature samples.
• Sleep mode when the ADC is sampling and direct memory access (DMA) is moving the results to internal memory.
• Active mode when the CPU is processing data.

You can also incorporate power monitoring and optimization at the application level. Let’s assume that you are measuring the temperature every few seconds and sending the measured values every few minutes. Along with this, you can add a battery monitor using the on-chip ADC peripheral to monitor the battery voltage level. When the battery voltage drops below the monitor threshold, the application can reduce the frequency of temperature sampling and communication over the network. Better still, the application can send a message to the host indicating a low battery level so that the user can change or charge the battery.

Tools that help you save power

Ultra-low power MSP430™and MSP432™ MCUs enable you to design your application power consumption to within your desired limits. Additionally, MSP430™ and MSP432 MCUs offer useful tools to optimize your application for ultra-low power operation.

Both new and advanced embedded system developers may find these tools valuable:

• The ultra-low power (ULP) advisor is a tool for guiding developers to write more efficient code in order to fully utilize the unique low-power modes and features of MSP430 and MSP432 MCUs.
• EnergyTrace™ software for MSP430 and MSP432 MCUs is an energy-based code analysis tool that measures and displays the application’s energy profile and helps optimize for ultra-low-power consumption.

Here are the key takeaways from the sixth blog post of this series:

• MCU low-power modes help you reduce overall device current consumption by gating clock and power to modules that are not required to be functioning at all times during application operation.
• Duty-cycle the different MCU peripherals according to your needs and power budget, and make use of the MCU power modes accordingly to optimize the overall energy consumption of your smart thermostat.
• Leverage MSP430™ and MSP432™ power- and energy-optimization tools to fully utilize its low-power features.

Stay tuned for the final part of this blog series.

Additional resources

• For more guidelines about calculating overall device power consumption and estimating the battery life of a specific battery, download the white paper, “Benchmarking MCU power consumption for ultra-low-power applications.”

Xavier Pedrosa, a hard-working farmer whose family has lived near Barcelona since the Middle Ages, never had much need for technology. In an age when many farmers have gone high tech, he still favors traditional, organic methods to grow cereal crops, olives and vegetables.

Then one day, his daughter, Rut, met Joaquim Oller. Joaquim was about to begin his doctorate in telecommunications engineering at the Polytechnic University of Catalonia (UPC) and Rut was about to begin her doctorate in marine biogeochemistry at the University of Barcelona.

Their friendship blossomed into romance, and within a few years the traditional farmer had an idea-generating, high-tech whiz for a son-in-law.

“Rut and her family are like magicians on the farm, while I don’t know anything about farming,” said Joaquim, who now lives with Rut in Cape Cod, Mass. “But I do enjoy making life easier for people.”

Joaquim began designing a device that can tell Xavier when his crops need water or when the soil is wet or dry enough to work it. His design, however, faced several challenges: It had to be easy to use, inexpensive, and not rely on batteries or solar panels in order to preserve the farm’s commitment to green operations.

Joaquim created a simple device featuring our MSP430™ microcontroller (MCU) to help Xavier monitor the level of moisture in the soil he farms. A battery-free sensor placed in the soil connects wirelessly to a smartphone app and provides information that helps him know when to plow and when to irrigate.

“I said to Rut, ‘Is this something your father would use?’ She told me, ‘Among all your silly ideas, this is a good one.’”

Early love of technology

Joaquim began his love affair with technology as a child at his home in a village near the ancient city of Girona, about 100 kilometers northeast of Barcelona. Founded in 79 BC, Girona’s rulers have included Romans and Visigoths. Charlemagne captured it in 785. In more modern times, the city’s cathedrals, winding alleys and plazas have played a leading role in the blockbuster show Game of Thrones.

But despite all the history, Joaquim’s childhood was focused on computers. He began programming at 10, excelled in school and earned a bachelor’s degree in computer engineering and a master’s degree and doctorate in telecommunications engineering from UPC, which is a science hub in Southern Europe.

As a professional, Joaquim has worked on projects for many companies, universities and clients through the years. He has deployed radio-frequency identification (RFID) payment technology for public-transportation systems, biometric readers, wake-up radio systems, sub-GHz nodes, agriculture designs, capacitive user interfaces and a host of other projects.

As a researcher, he developed expertise in low-power microcontrollers and Bluetooth® low energy (BLE) technology. As part of the Wireless Networks Group research lab at UPC, he conducted the technical performance evaluation of a scientific paper that became a primary BLE resource. The paper, which references our technology extensively, has been downloaded about 50,000 times.

As a member of TI’s engineer-to-engineer (E2E™) community, Joaquim has engaged in nearly 1,000 online conversations to help others understand technology and overcome design challenges. His E2E username is Kazola. E2E, which launched in 2008, is a technical support community of nearly 300,000 engineers and TI experts spanning more than 200 countries who collaborate by asking and answering technical questions, sharing knowledge, exploring new ideas and solving problems.

Technology as a farm tool

Xavier’s farm, anchored by a home his ancestors originally built some 1,300 years ago, is at the base of the jagged Monserrat range, which is widely known for an ancient mountaintop Benedictine monastery. In addition to crops, he has begun raising goats for milk and cheese and to help control brush as part of a European project to prevent wildfires.

In such an idyllic place, technology is being introduced gradually. But if Joaquim’s idea saves money, time and water, Xavier may find other uses for it in the future.

“Technology has to be employed as an additional tool and not as a substitute one,” Xavier said. “The main tool for successful farming is and will always be human knowledge and intuition.”

Like most people on a wicked hot sunny day, all I can think about is how to beat the heat, which more often than not involves hiding in an air-conditioned room in my home.  Later – on almost always a cooler day, with open windows – I’ll receive the electric bill and ask myself whether that temporary comfort was really worth it.

The air conditioner can only achieve a pyrrhic victory against the sun and its heat.  As an engineer, I took this as a problem that needed solving.  My solution is a simple one: if you can’t beat ’em, join ’em.  Rather than consume large amounts of expensive utility electricity, it would be better if solar panels on the roof powered the air conditioner.  Luckily for me I am not the first to think of this, and with costs of solar power nearly on parity with traditional energy sources, this goal is starting to become a reality for everyone.

While a lot of attention goes to the photovoltaic panel itself, the rest of the solar power ecosystem needs to advance as well.  The power electronics are but one critical aspect.  Since photovoltaic panels generate a DC voltage but the electricity transmission and distribution system is in AC, a power inverter is required.

To meet the cost goals of solar power systems, the U.S. Department of Energy came up with the following specifications as part of the Sunshot Initiative:

• Conversion efficiency >98%
• Service life >25 years
• Power density >100W/in³
• System cost <$0.10/W (utility); <$0.125/W (commercial); <$0.15/w (residential) (This includes the lifetime cost of the power-electronic device, including the initial capital cost and operation and maintenance (O&M) costs over its service life)

The Sunshot Initiative targets are aggressive; to meet them requires not just the optimization of the core, but careful consideration of every part.  One place where careful design can lead to a large impact is the isolation boundary.  The inverter’s connection between a low-voltage DC and dangerous high-voltage AC requires galvanic isolation, which can lead to a situation where a power field-effect transistor (FET) or insulated-gate bipolar transistor (IGBT) is on the opposite side of the isolation barrier from the controller generating the gate signal.  Reinforced isolated gate drivers such as TI’s UCC21520 are great because they combine multiple functions, passing signals across an isolation boundary and converting logic gate signals into an actual gate drive, from several devices to just one.

The UCC21520 improves on these integration benefits by having leading performance for propagation delay and delay matching between the high and low side.  This reduces losses associated with the switch since it turns on faster and also reduces the required dead time, which is when the higher-loss body diode conducts.  These parameters are also less dependent on VDD, so you can relax the design margin for voltage tolerances in the rest of the system, as the bench data in Figure 1 shows.  Figure 1 also highlights that the UCC21520 performs significantly more linearly over VDD than its competition.

Figure 1: TI’s UCC21520 propagation rise/fall delay with respect to VDD vs. a competitor

Another device that needs to cross the isolation boundary is the auxiliary power supply.  To ensure that the solar inverter is running and “smart” – regardless of the state of the AC utility/load or photovoltaic (PV) panel – requires an isolated power supply to provide bias power for the inverter.  Since this stage crosses the isolation boundary, it requires components that cross it as well.  Primary-side regulation (PSR), where the output is regulated from an auxiliary winding that is grounded with respect to the primary-side controller rather than an optocoupler, is a great way to reduce components and cost.  PSR also has the added benefit of increasing lifetime as well, since it eliminates a notorious point of failure during surge voltages.  Primary-side flyback controllers such as the UCC28700 maximize performance and efficiency of the control scheme by implementing advanced algorithms with minimal external circuitry.  The UCC28910 expands these advantages by integrating the 700V power FET and controller into a single device, further reducing the size of the bias power supply.

TI solutions can help make cheap, reliable solar energy – enough to power an air conditioner in the middle of a heat wave.

Additional resources

• Download the white paper, “Isolation in solar power converters: Understanding the IEC62109-1 safety standard”
• Discover TI’s smart grid solutions, including solar energy
• Learn more about PSR flyback converters
• Start designing a 25W auxiliary power supply using the UCC28700 PSR flyback controller with the 25W 88% Efficiency Multiple Isolated Output Auxiliary Supply Reference Design for AC/DC Power Supply

Capacitive loads can cause stability problems in operational amplifier (op amp) circuits, resulting in large overshoots, ringing, long settling times – and in severe cases, sustained oscillations. These issues occur because the capacitive load interacts with the op amp output impedance, forming an additional pole in the open-loop gain (Aol) response that reduces the loop-gain (Aol*β) phase margin below acceptable levels.

Many resources present basic stability theory in great detail, including TI Precision Labs’ videos on op amp stability. There are different compensation circuits which allow the op amp to remain stable while driving the capacitive load. In this blog post, I’ll review three common compensation circuits that can be designed and tested using the do-it-yourself amplifier evaluation module (DIYAMP-EVM).

Isolation resistor – R_ISO

The most common and easiest-to-design method places an isolation resistor (R_ISO) in series with the capacitive load. The isolation resistor adds a zero to the Aol*β transfer function, which cancels the phase shift from the pole and returns the rate of closure (ROC) to 20dB/decade. As shown in Figure 1, this compensation circuit produces stable results with phase margins greater than 60 degrees by following the design steps in the TI Precision Labs videos and placing the zero at or above the frequency where the loaded Aol curve is equal to 20dB. Moving the zero higher in frequency lowers the phase margin to achieve a more critically damped response. The main disadvantage to this compensation method is that there will be a voltage drop across R_ISO, which reduces the DC accuracy of the circuit when driving a load.

Figure 1: R_ISO capacitive load compensation circuit and open-loop results

R_ISO + DFB circuit

A common solution to maintain DC accuracy while stabilizing the load is to use the R_ISO plus dual-feedback (R_ISO + DFB) circuit. As the name suggests, this compensation circuit has two feedback paths. There is a DC feedback path through RF that regulates the voltage at the load and an AC feedback path through CF, which makes the circuit act like the R_ISO circuit at high frequencies to stabilize the capacitive load. Be sure to follow the guidelines in the TI Precision Labs videos for setting the feedback components to achieve proper operation.

Figure 2 shows the open-loop results for the R_ISO + DFB circuit. While this circuit restores the DC accuracy lost while using the R_ISO circuit, the R_ISO + DFB circuit has slower settling times than the original R_ISO circuit.

Figure 2: R_ISO + DFB compensation circuit and open-loop results

R_ISO + DFB + RFx circuit

Another compensation method that maintains DC accuracy but offers improved transient response is the R_ISO + DFB + RFx circuit. This circuit is basically the same as the R_ISO + DFB circuit with an additional resistor, RFx, in series with CF. The addition of RFx will cause the circuit noise gain (1/β) to increase and flatten off at higher frequencies to a magnitude of 1 + RFx/RF. This increase in noise gain can be helpful to compensate difficult capacitive loads, lower the loop-gain crossover frequency, adjust the phase margin and shape the closed-loop output impedance. For this circuit to be stable, the 1/β response must flatten off to the high-frequency level before it intersects with the Aol curve to achieve a 20dB/decade ROC. Figure 3 shows the Riso + DFB + RFx circuit and the open-loop results.

Figure 3: R_ISO + DFB + RFx compensation circuit and open-loop results

Figure 4 compares the transient results to a small-signal step input at both the op amp output (Vo) and the circuit output (V_OUT). The R_ISO circuit displays a typical overdamped response with a moderate settling time. The R_ISO + DFB circuit output has a rounded single-lobe overshoot and long settling tail resulting in slower settling times than the R_ISO circuit. The R_ISO + DFB + RFx output has a sharp single-lobe overshoot but settles very quickly to the final output value with a similar settling time to the original R_ISO circuit.

While the overshoot spike at the op amp output in the R_ISO + DFB + RFx circuit (Vo_RFx) looks troubling, the behavior is a result of the complex nature of this circuit’s transfer function and how the placement of the poles and zeros affects the transient response. Don’t worry; the circuit is stable. The overshoot is aperiodic and not followed by the substantial ringing that would normally be associated with this level of overshoot if the circuit was unstable.

Figure 4: Comparison of small-signal step responses for the three compensation circuits

Figure 5 compares the total output noise of the three circuits. The R_ISO circuit has the lowest noise, followed by the R_ISO + DFB circuit and then the R_ISO + DFB + RFx circuit. The R_ISO + DFB + RFx circuit has the highest noise because of the increase in noise gain at higher frequencies. Circuits with higher ratios of RFx/RF will have more noise than those with smaller ratios.

Figure 5: Total output-noise comparison for the three compensation circuits

You can evaluate all three of these circuits using the R_ISO + DFB circuit in the DIYAMP-EVM shown in Figure 6. Table 1 lists the component configurations to create each of the three circuits using the DIYAMP-EVM R_ISO + DFB circuit.

Figure 6: R_ISO + DFB circuit in the DIYAMP-EVM

Table 1: R_ISO + DFB DIYAMP-EVM circuit component configurations

I hope that after reading this post, you’ll feel confident doing your own stability analysis and compensation using the DIYAMP-EVM.

Additional resources

• Check out the DIYAMP-SOIC-EVM User’s Guide.
• Download the “Analog Engineer’s Pocket Reference.”
• Learn more about the TLV9062 op amp.

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When I was six years old, I got a remote-control car as a present from my dad. With just a click of a button, I was able to control the car and it was running all around my home. One day, my brother and I suddenly got into a fight and broke the remote-control car into pieces. I was curious to look inside and see how it worked. I learned that remote-control cars used batteries as their main source of energy. I went to my father and asked, “Do all cars operate with batteries?” My father laughed and said, “No, it is impossible to power a car with a battery; only toy cars can operate with batteries. Cars on the road operate with fuel.”

Whenever I remember this story I am quite amazed, because now I work on systems for electric vehicles (EVs), and I know that it’s 100% possible to power a car with batteries. The main source of energy for hybrid EVs (HEVs) and EVs is the battery; efficient control of the battery implies an efficient battery-monitoring system.

The battery-monitoring system is mainly used to estimate state of health (SOH) and state of charge (SOC). In order to obtain detailed information about SOH and SOC, integrating accurate sensors into the battery-monitoring system is important. For a typical battery, current, voltage and temperature sensors measure the following parameters, while also protecting the battery from damage:

• The current flowing into (when charging) or out of (when discharging) the battery.
• The pack voltage.
• The individual cell voltages.
• The temperature of the cells.

Figure 1 shows the location of current sensors in a block diagram of a battery-control unit.

Figure 1: Current-sensor location in battery-control unit

When the battery is the main source of energy for systems in HEVs/EVs, it is essential to have information about its charging and discharging cycles. Current sensors are the main source of information for charging and discharging cycle information by reporting the status of battery SOH to the battery management system. They may be located onboard or externally. With the increase of battery capacities in HEVs/EVs, the requirements on higher current ranges are increasing. Here are the main requirements for a typical current sensor in HEVs/EVs:

• A current range from milliamps to kiloamps for example, 2000A to 2000A, -1200A to 1200A and -500A to 500A. Higher current ranges are required in order to accommodate larger battery capacities, monitor dynamics of the load such as peak current detection (shorts to battery/shorts to ground), and satisfy initial startup/torque demands.
• Higher Bandwidth. Higher bandwidths are required in order to monitor dynamics of the load or respond to fault states. Peak currents are in the kiloamp range and last from milliseconds to few seconds.
• High accuracy. A battery-management system’s ability to accurately measure parameters such as pack voltage, charging/discharging current, individual cell voltages, battery disconnection in abnormal conditions, charge stored by each cell in a stack, operational status of system components for assistance with functional safety, SOC, SOH and state of function (SOF) all depend on the accuracy of the sensor inputs. Accuracy of the current sensor is important, especially at lower currents, to take decisions in a very prompt way to increase system efficiency. Accuracy is normally specified separately for lower and higher currents.
• Temperature and linearity compensation. Temperature and linearity are critical factors in current sensors because of their dependency on temperature. Temperature dependency on the system results in poor accuracy. Maintaining the same accuracy over the entire temperature range is essential. Satisfying this requirement requires a temperature and compensation algorithm.

Battery current sensors can be realized in two ways – shunt and magnetic – as shown in Figure 2.

Figure 2: Physics of current sensing

Both shunt and magnetic technologies have their own advantages and disadvantages.

Shunt-based current sensors

A finite amount of current passes through a shunt resistor; an analog front end (AFE) amplifier measures the voltage drop. With advancements in low-value precision shunts and huge improvements in AFE circuits, shunt technology has been widely adapted for measuring currents in HEVs/EVs. For example, shunts from Vishay offer very low resistance values (50μΩ, 100μΩ, 125μΩ and 500μΩ). With these lower-value shunts, the voltage drop across the shunt is very small. Now the challenge is to measure that tiny voltage drop. TI’s automotive current sensors provide complete solutions for these kinds of systems. 

Nonisolated shunts

The Automotive Shunt-Based ±500A Precision Current Sensing Reference Design offers battery current sensor solutions, providing excellent accuracy and linearity over a grade 1 (-40°C to +125°C) temperature range. Figure 3 shows how the reference design addresses the requirements.

Figure 3: Requirements and TI response

Offering such performance characteristics over the automotive grade 1 temperature range (-40°C to +125°C) makes this reference design successful in battery current-sensing applications. Signal conditioners like the INA240-Q1 and PGA400-Q1 enable more efficient HEV/EV battery current sensing. Figure 4 shows a block diagram of the main components in a battery current sensor.

Figure 4: Nonisolated current sensor block diagram

Isolated shunts

Devices like the AMC1301 and ISOW7821 offer isolated shunt current-sense measurement. By providing excellent isolation between the hot and cold sides, this system helps with battery sensing reliability. As shown in Figure 5, the AMC1301 provides current signal isolation and the ISOW7821 provides power isolation. The PGA400-Q1 completely eliminates the offset and gain errors.

Figure 5: Isolated current sensor block diagram 

One of the biggest reasons for the evolution of HEVs/EVs is emerging battery technologies. Battery performance, lifetime, safety and reliability play a vital role in HEVs/EVs. A battery current sensor and its accuracy over a wider range are extremely important in order to achieve the required parameters. TI’s battery current-sensing portfolio enables you to achieve these specifications easily and simply.

Additional resources

• Learn more about technology advancements in HEV/EV.
• Read these blog posts:
  •  “Interconnecting automotive 48V and 12V rails in dual-battery systems.”
  • “Learn how to keep your car battery steady and running longer.”

For most people, listening to frequency modulation (FM) radio is a part of their life. But what is FM exactly? It’s simply a popular modulation scheme used to embed information on a high-frequency radio carrier. The hardware requirements for implementing FM are low. Nevertheless, there are a couple of FM implementation techniques, each with different characteristics.

FM implementation

In general, you can implement FM in either an analog or digital way. Regardless of technique, you will need a frequency synthesizer (Figure 1) to generate the high-frequency radio carrier.

Figure 1: Frequency synthesizer simplified block diagram

A frequency synthesizer is a closed-loop system that consists of a clean and stable low-frequency reference clock (RefClk), a phase-locked loop (PLL) chip, a loop filter that defines the bandwidth of the closed loop and a high-frequency voltage-controlled oscillator (VCO). The VCO will track the RefClk. Its frequency is equal to N x RefClk, where N is a number greater than or equal to 1. The loop filter response is low-pass to the RefClk and the PLL. That is, the RefClk and PLL outputs will be low-pass-filtered before going to the VCO. The loop filter response to the VCO, however, is high-pass-filtered.

Depending on the FM implementation technique, the modulation signal (the information that you want to embed in the high-frequency carrier) is applied to modulate the RefClk, VCO or PLL. The result of the modulation is that the carrier frequency will shift continuously. The amount of frequency shift is called frequency deviation. Figure 2 depicts a FM signal in time domain and modulation domain.

Figure 2: FM modulation

FM implementation – analog technique

In the analog approach, the modulation signal is applied to either the RefClk or VCO. The advantages of this method are that the hardware is very simple and easy to implement. The drawbacks of this method are that performance is not consistent, or requires special care to make it consistent across different VCOs and modulation frequencies.

For example, let’s assume that RefClk = 20MHz and VCO = 480MHz. Since VCO = N x RefClk, N = 24. If a 1Vpp modulation signal is applied to the RefClk and produces a ±100Hz frequency deviation at the output of the RefClk, then the frequency deviation at the output of the VCO becomes ±2.4kHz. To achieve the same VCO frequency deviation for VCO = 960MHz, you must adjust the strength of the modulation signal, as N has become 48. Unfortunately, this adjustment may not necessarily be linear. In other words, a 0.5Vpp modulation signal may not return a ±50Hz frequency deviation. This is true even if the modulation signal is applied to the VCO. The VCO gain (Kvco) changes over the VCO frequency. Kvco refers to how the VCO frequency would change vs. the control voltage. If Kvco is not linear across the entire VCO operation range, then the same modulation signal strength applied to the VCO will result in different frequency deviations when the VCO frequency changes.

Furthermore, the loop bandwidth (LBW) will determine the usable modulation frequency. If the modulation is applied to the RefClk, the maximum modulation frequency will be less than the LBW because the loop filter is low-pass-filtered to the RefClk. In fact, the minimum modulation frequency has to be greater than the LBW if the modulation is applied to the VCO. In some use cases requiring a flat response, the modulation signal will be applied to both the RefClk and VCO simultaneously. Figure 3 shows the loop filter response to the RefClk and VCO.

Figure 3: Analog implementation technique

FM implementation – digital technique

You can overcome the shortcomings of the analog technique by digitally synthesizing the FM modulation through the PLL. If VCO = N x RefClk, by continuously changing the N value in a precise manner, you can synthesize the modulation domain view waveform, as shown in Figure 2. The advantages of this method are that the frequency deviation does not depend on Kvco and the VCO frequency anymore. In addition, if the LBW is wide enough, the modulation frequency response will be flat. The trade-off is that this method requires higher digital processing power from the logics because the PLL requires continuous programming. Plus, the synthesizing rate (or sampling rate) must be higher than the modulation frequency in order to reduce the sidebands created from sampling. For more details on this method, see the application report, “Frequency Shift Keying with LMX2571.” Figure 4 shows how the desired waveform is sampled and synthesized with LMX2571.

Figure 4: Digital implementation technique

Which technique is better?

In analog FM radio broadcasting system, the analog technique will be better. It is because the carrier is usually fixed at a particular frequency. The shortcomings of this technique do not happen at all.

In applications with multiple carrier channels, for example, 2-way radio, the digital technique will be superior to the analog technique.

In general, both techniques have their pros and cons. The adoption will depend on the application needs, system capability, performance index, and cost.

Additional resources

• Read the blog post, “Timing is Everything: Understanding PLL loop filter response.”
• Check out the LMX2594 Datasheet
• Explore TI’s entire PLL/synthesizer portfolio.

A typical power source for a portable device like a smartphone is a single lithium-ion battery. With the development of silicon anode material and maximizing the battery energy as much as possible, the minimum operating voltage in a smartphone application for example, is usually lower, for instance 2.7V.

Some load configurations, such as Wi-Fi® modules, embedded multimedia cards (eMMC) and Secure Digital (SD) cards, require a regulated voltage greater than 2.7V. The low-dropout regulators (LDOs) integrated into the power-management integrated circuit (PMIC) supply these power rails. The LDOs’ input (VLDO_IN) must be slightly higher than the highest LDO’s output voltage. Therefore, if VLDO_IN ends up in the middle of the lithium-ion battery’s operating range, a boost regulator is necessary between the battery and the LDOs’ input end to guarantee that the LDOs’ input is higher than their highest regulated voltage.

If the battery is in the well state of charge and the voltage is higher than the required minimum system voltage, the load configurations don’t need a boost function, but just directly bypass the battery voltage to the LDO’s input.

Figure 1 illustrates the power system using the boost bypass as a pre-regulator in a smartphone application.

Figure 1: Boost with bypass for the PMIC pre-regulator

TI developed the TPS61280A, TPS61281A and TPS61282A PMIC family especially for the pre-regulator of the smartphone.  It operates in a low-ohmic, high-efficient bypass mode when the battery voltage is higher than the minimum required voltage of the LDOs’ input. If the battery’s voltage becomes lower than the required minimum voltage, the device seamlessly transitions into boost mode, as shown in Figures 2 and 3.

Figure 2: TPS618xA boost/bypass connection

Figure 3: TPS618xA output voltage regulation

Sweeping the input voltage of the TPS6128xA, with the bench measurement of TPS6128xA conditions at V_{OUT_Boost} = 3.4V, the output voltage follows the input voltage, with a gap around 70mV at a 1.5A load in bypass mode, which is caused by the current flowing through the bypass field-effect transistor (FET) (M3).

During input voltage ramp down, when the input voltage crosses the boost/bypass threshold (3.4V in this case), the TPS61280A enters into boost mode, with around 100mV of undershoot at the output of the TPS61280A.

When ramping up the input voltage, the TPS61280A enters into bypass mode as long as the output voltage is 2% higher than the 3.4V threshold, and the boost-to-bypass has no undershoot (or overshoot) with very smooth transition.

As an example, consider bench measurements with the conditions at load = 1.5A, output capacitance = 16µF (effective), and sweeping the input voltage from 3.3V to 3.7V. In this case, the input voltage is larger than the desired target voltage and the output of the TPS61280A follows the input voltage with a drop voltage of the bypass FET. The output voltage keeps at the target value when the input voltage sweeps below the target value as shown in Figure 4.

Figure 4: TPS6180A output voltage regulation

With a smooth transition between boost and bypass as well as the high efficiency in either boost or bypass mode, the TPS6128xA enables the use of the full battery capacity. You can overcome a high battery cut-off voltage originated by powered components with a high minimum input voltage and silicon anode discharge battery chemistries. The device buffers high current pulses forcing the system into shutdown, seamlessly transitioning between boost and bypass mode.

The pre-regulator benefits with extending the battery on-time and this has a significant impact on battery on-time and translates into either a longer use time or a better user experience at an equal battery capacity, or into reduced battery costs at similar use times.

The TPS61281A and TPS61282A have a fixed current-limit threshold as well as a default voltage value where the transition between boost and bypass takes place. The TPS61280A is a fully programmable device via I²C (two-wire interface). The TPS61280A gives you a high level of flexibility to tailor the device to your own system needs. Table 1 summarizes the key parameters of the TPS6128xA device family.

Table 1: TPS6128x family device overview

TI’s TPS61280A, TPS61281A or TPS61282A PMIC used as a pre-regulator extends the battery run time and overcomes input current and input voltage limitations of the system being powered. Offered in a 16-pin chip-scale package (CSP), the PMIC provides a very small total solution footprint (<20mm²) with minimal external inductor and input capacitors. Get more information on TI’s family of boost converters with integrated switches for Li-ion battery-powered applications.

In the first installment of this series, I stated that there are many reasons for switched-mode power supply (SMPS) instability, only one of which is that the control loop has insufficient gain or phase margin. In this installment, I will offer some tips about identifying and curing subharmonic oscillations in peak current mode (PCM)-controlled SMPS systems and talk briefly about input-filter oscillations. 

Subharmonic oscillations

There is a well-known, inherent instability in continuous conduction mode (CCM) PCM control loops when they operate at duty cycles greater than 50%, as shown in Figure 1. Discontinuous conduction mode (DCM), transition mode (TM), average current mode (ACM) and voltage mode-controlled (VMC) systems are not susceptible to this type of instability. But be careful – because DCM, TM, ACM and VMC systems often use PCM control when they operate in current limit.

Figure 1: Subharmonic oscillations

Diagnosis and solution

Subharmonic oscillations appear as large changes of duty cycle from cycle to cycle. They usually persist because the average duty cycle remains greater than 50%, but they can appear transiently if a load change causes the controller to run at a more than 50% duty cycle for a few cycles. It’s also worth noting that without slope compensation, current perturbations take longer and longer to die out as the duty cycle increases towards 50%. Here is a short list of the behaviors you might see.

• Does the problem disappear at duty cycles less than 50%? If so, the solution is to correct the amount of slope compensation.
• Increase the inductor value so that you need a slower slope compensation ramp.
• Reduce the loop bandwidth – loop bandwidths that are more than about one-fourth of the switching frequency can become unstable if subharmonic oscillations become established.
• It may be possible in some cases to change the transformer turns ratio or the operating range of the SMPS so that it never exceeds a 50% duty cycle.

Input-filter oscillations

Most power supplies present a constant power load to their inputs and therefore have a negative incremental input resistance. This means that the input current will decrease as the input voltage increases. In an offline power factor correction (PFC) stage, the current control loop forces the system to emulate a positive resistance at line frequencies so that the input current follows the sinusoidal shape of the input voltage. But the negative input resistance behavior is present at frequencies beyond the control loop crossover.

DC/DC and offline AC/DC converters will normally have some form of input filter like that shown in Figure 2. This filter is necessary to meet conducted electromagnetic interference (EMI) requirements but it can oscillate under some circumstances if not designed correctly. This topic has been widely discussed in literature, but the summary rule is simple enough: the output impedance of the filter must be less than the input impedance of the converter at all frequencies.

Figure 2: Typical AC/DC converter input filter, with the SMPS input impedance in green, the undamped filter output impedance in red and the damped filter output impedance in blue

Diagnosis and solution

The simplest way to identify an input-filter oscillation is to remove the input filter by short-circuiting the input-filter inductors. The filter will normally oscillate at one or other of the resonances of the filter. These resonances are normally in the range between 1kHz to 10kHz depending on the filter design. Curing input-filter oscillations requires modifying the input filter to reduce its output impedance while maintaining its effectiveness. Here are the two things to try:

• Change the L and C values to reduce the impedance.
• Add damping resistors to reduce the filter impedance at its resonant frequencies (also called the Q factor). You can compare the red (undamped) and blue (damped) traces in Figure 2.

So far, I have discussed classic feedback loop instability, subharmonic oscillations and input-filter oscillations. In the next installment, I’ll look at oscillations caused by remote sensing connections.

The rapid growth of USB Type-C enables a unified interface to deliver data, video, audio and power through a single connector. USB Type-C™ not only supports the latest USB data rate of 10Gbps, it also enables up to 100W power delivery through the same cable carrying the data traffic simultaneously. Type-C has reversible plug orientation; it also allows host and device to perform data role swap and power role swap providing ease of use and flexibility. In addition, USB Type-C can also deliver high resolution video protocol over Type-C interface, eliminating different cable and connector needs thus greatly simplify and enhance user experience. Billions of devices will use the USB Type-C interface to connect in the coming years.

You can integrate many popular protocols such as DisplayPort™, Thunderbolt and High-Definition Multimedia Interface (HDMI) into USB Type-C through Alternate Mode. Ensuring signal integrity for these various protocols over four high-speed lanes represents a challenge for designers, however.

The TUSB544 is the industry’s first USB Type-C Alternate Mode multiprotocol bidirectional linear redriver supporting USB 3.1 at 5Gbps and DisplayPort 1.4 at 8.1Gbps. TUSB544 resolves signal-integrity issues at the USB Type-C connector to compensate channel loss for USB hosts and devices, thus output clean Type-C signals. For DisplayPort source or sink devices, placing the TUSB544 inside the USB Type-C cable enables the use of longer cables and improves signal quality.

There are many use cases for the TUSB544 in a system. Some central processing units (CPUs) and mobile processors have an integrated USB Type-C multiplexer in the processor; the CPU will multiplex 4 DisplayPort™ lanes and 2 USB lanes into 4 high speed output, each lane can be individually configured as USB or DisplayPort. If you place the TUSB544 near the connector – on either the transmitter or receiver side – you can configure the four USB Type-C lanes as negotiated by the source and sink devices and redrive the signal in either direction according to your protocol needs. For example, the signal over USB Type-C can be USB only, four-lane DisplayPort only, or two-lane USB and two-lane DisplayPort. Figure 1 shows an end-to-end USB Type-C solution using the TUSB544 as a redriver.

Figure 1: Enabling better signal quality with the TUSB544 as an end-to-end solution

The TUSB544 linear redriver can reduce design complexity and provide device placement flexibility in your system. The device particularly benefits DisplayPort applications because it is transparent to link training, establishing a better channel with a minimal bit error rate for clear video displays.

Another use case for the TUSB544 is to create active cable for extending USB Type-C cable lengths because you can place the redriver inside the cable to compensate for cable loss. Figure 2 shows the eye diagram of a 3m USB Type-C cable without any redriver. The red circles indicate that the eye diagram is failing the eye mask.

Figure 2: A 3m USB Type-C cable failing the eye diagram

By applying the TUSB544 onUSB Type-C high-speed lanes and the TUSB211 on USB 2.0 lanes, you can improve the signal quality for all USB 3.1, DisplayPort and USB 2.0 signals at the USB Type-C interface. Figure 3 shows a USB Type-C active cable solution with the TUSB544 and TUSB211.

Figure 3: USB Type-C active cable solution with the TUSB544 and TUSB211

After adding the TUSB544, the eye diagram passes, with excellent performance for a 3m USB Type-C cable, as illustrated in Figure 4.

Figure 4: A passing eye diagram with the TUSB544 inside a 3m USB Type-C cable

To enable even longer USB Type-C cables, you can place two TUSB544 redrivers at both ends of the cable to provide enough channel-loss compensation. You will see excellent performance as well with a 5m USB Type-C cable that has a TUSB544 placed at each end, as shown in Figure 5.

Figure 5: A passing eye diagram of a 5m USB Type-C cable with two TUSB544 devices at each end

The TUSB544 offers flexibility when redriving USB Type-C signals for USB and DisplayPort applications to deliver clean and excellent signal quality. It helps systems pass compliance tests, enables better interoperability and improves USB Type-C system performance.

Additional resources

• Download the TUSB544 data sheet.
• Read the white paper: “Alternate Mode for USB Type-C: Going Beyond USB”.
• Watch the video: “Strengthening signal chain through signal conditioners”.

In our ongoing series, ‘One to Watch,’ we profile TIers who are making a difference through innovation or citizenship.

It is hard to choose a power topology for a wireless design when there are so many power-management integrated circuits (IC) out there. It may be hard to tell where to start, so I always start at the selection basics. The application of the design and onboard devices will determine the different voltage rails needed for operation.

Consider a smart lock or a heating, ventilation and air conditioning (HVAC) damper control system. This control system usually consists of a higher-voltage motor driver than the rest of the board The lower-voltage rail powers the MCU, radio and other sensing components.

A power management IC will make creating the different voltage rails easier. In the smart damper system, you have two ways to create two different voltage rails. The first option is to create the 3V rail with the batteries and then boost the voltage up to 6V for the motor driver. The second option is to have the batteries supply the 6V for the motor driver and then step the voltage down to 3V for the rest of the system.

The next step in the design is to choose a power topology. Three main power topologies are possible for smart damper applications: low dropout (LDO) regulator, step-down converter (also known as buck) and boost (see Figure 1). The LDO and buck implementations are not event-dependent, meaning that the two topologies use the same amount of energy regardless of how many damper louver movements occur in a day. The LDO and buck will drop the voltage down to generate the lower-voltage power rail to run the microcontroller (MCU), and the higher-voltage components run off the higher battery voltage. The boost is event-dependent because each damper adjustment event must boost up the voltage from 3V to 6V for motor and light-emitting diode (LED) operation.

Figure 1: Power topologies

I chose a buck configuration because the LDO has ground leakage current, whereas the buck has zero ground leakage current and therefore more efficiency.

For more analysis into power topologies for smart lock and HVAC damper systems, check out our reference design guides. The Smart Lock Reference Design Enabling 5+ Years Battery Life on 4x AA Batteries and Smart Damper Control Reference Design With Pressure, Humidity and Temperature Sensing both include more in-depth analysis into the various power topology choices.

Skipping ahead to the part choice, I chose the TPS62745 step-down converter because of the extra benefits it offers for low-power designs. This device has select lines that enable users to select the output voltage, and therefore removes the need for a feedback resistor, thus slightly decreasing bill of materials (BOM) cost. The TPS62745 can dynamically enable or disable the battery voltage check with the use of an enable pin and an external resistor voltage divider. Energy is conserved because the voltage divider is only enabled when a battery voltage check is necessary. The rest of the time, the divider circuitry does not use energy and is not connected. The TPS62745 is also efficient at extremely light loads; it is 85% efficient at 10μA. Efficiency is key because the smart damper systems are asleep much of the time.

Figure 2: Smart damper reference design block diagram

With your power topology and management IC chosen, all you have to do now is select the parts and sensors that connect to each of the power rails. TI has a wide selection of power-management ICs and different flavors of MCUs to suit your system needs.

Additional resources

• Read these E2E™ Community blog posts:
  • “How to approach a power-supply design – part 1.”
  • “How to approach a power-supply design – part 2.”
  • “Increasing battery life in IoT devices.”
• Design your power stage with Power Stage Designer.
• Take a look at the “Power Topologies Handbook” and “Power Topologies Quick Reference Guide.”

For nearly four decades, the Texas Instruments Power Supply Design Seminar has brought in-person training from power-supply experts directly to customers.

For each seminar, we carefully choose topics under guidelines set by Bob Mammano. Ultimately, each topic selected must be useful, educational and interesting. We take pride in the fact that the seminar is not a sales pitch, but truly an educational experience.

Our most recent 2016/2017 seminar was the 22nd edition (SEM2200), which wrapped up earlier this year after touring the United States, China, India, Japan, Taiwan, Korea and Europe. Personally, I am thankful to have been a part of the seminar tour, and to have had the opportunity to meet so many fellow engineers in the power-supply community.

If you missed the last seminar tour, it’s not too late; we are now preserving the experience for you online through a new training site, with all of the 2016/2017 seminar material in one place. With a myTI login, you can access videos of each presentation and downloadable versions of the papers and presentation material. Each video has been studio recorded and is around 40 minutes in length.

SEM2200 includes seven topics covering a variety of power-supply-related issues, written by expert engineers with practical experience dealing with those topics. They are:

• “Design of a high-frequency series capacitor buck converter” by Pradeep Shenoy. In this paper, Shenoy introduces the series capacitor buck-converter topology and discusses how it can significantly reduce the size of point-of-load (POL) voltage regulators. He also covers the limitations of conventional high-frequency buck converters and how the series capacitor buck converter overcomes these challenges.

• “Flyback transformer design considerations of efficiency and EMI” by Bernard Keogh and Isaac Cohen. The flyback converter is widely used in AC/DC power supplies due to its simplicity and wide operating range. In this topic, Keogh and Cohen focus on the importance of transformer design, since this single component has an enormous impact on converter efficiency and electromagnetic interference (EMI) performance.

• “Switch-mode power converter compensation made easy” by Bob Sheehan and Louis Diana. Compensating power supplies can be an arduous task for those not well versed in it. Sheehan and Diana break down the procedure into a step-by-step process that you can easily follow to compensate a power converter, while also explaining the theory of compensation and why it’s necessary.

• “Bidirectional DC/DC converter topology comparison and design” by Zhong Ye and Sanatan Rajagopalan. A bidirectional DC/DC converter is a key element of many new applications, such as automotive, server and renewable-energy systems. For this topic, Ye and Rajagopalan use a 48V/12V bidirectional converter as an example with which to revisit the hard-switching synchronous buck topology and compare it to a transition-mode totem-pole zero-voltage-switching (ZVS) topology.

• “SiC and GaN applied to high-frequency power” by John Rice and Rais Miftakhutdinov. Emerging wide-bandgap (WBG) silicon carbide (SiC) and gallium nitride (GaN) power devices are steadily gaining popularity in power electronics and have the potential to significantly increase a power converter’s efficiency and power density. In this paper, Rice and Miftakhutdinov examine important design issues when using WBG devices including drive technique, mitigating layout and packaging parasitics, high-frequency measurements, and simulations.

• “Under the hood of a noninverting buck-boost converter” by Vijay Choudhary, Timothy Hegarty and David Pace. When it comes to designing buck-boost converters, there is a huge gap between the simple inverting buck-boost converter in textbooks, which actually produce a negative output voltage, and real-world buck-boost applications that require a positive output. With this paper, Choudhary, Hegarty and Pace fill a gap in buck-boost literature by presenting various topologies used in noninverting buck-boost designs.

• “Design review of a 2-kW parallelable power-supply module” by Roberto Scibilia. In this paper, Scibilia steps through the design procedure of a real project that resulted in a prototype for a 2kW power-supply module. He covers the selection of the main power stages, including a continuous conduction mode (CCM) power-factor-correction circuit and a peak current mode-controlled isolated DC/DC resonant phase-shifted full-bridge converter with synchronous rectification.

I hope that you find the material from the seminar series truly useful, educational and interesting. Please share your questions and thoughts on the SEM2200 topics or the seminar in general by commenting on this post. Of course, as we are busy preparing for the next seminar series, we are very interested to hear what topics you would like to learn more about as well.

It’s no secret that when students get into the industry, they will need to have a basic understanding of IoT and how it works within embedded systems. Below we’ve compiled resources and materials to help you teach IoT in your courses using the TI LaunchPad™ development kit, ARM® Cortex®-M and TI's Real-Time Operating System.

Below we’ve included resources and tutorials that will give you a great low cost way to introduce the excitement of electronics and integrating wireless connectivity with IoT examples. Leave us a comment if you have any questions as you implement IoT into your embedded systems courses!

Watch our Facebook Live video from ASEE to learn more about teaching IoT with TI technology:

Commonly Asked Questions:

Does it cost anything to use an RTOS?

• Sometimes you have to pay to use an RTOS (real time operating system), but that doesn’t have to be the case anymore. TI-RTOS is free to use and easily accessible. There are many other free options like FreeRTOS or creating your own.

Why do you recommend teaching ARM in university courses?

• ARM is the standard for the industry, as such, knowledge of ARM can help students find employment. TI has a good ARM portfolio for learning, which can help make students attractive for job opportunities.

What is the total cost of your example in the video?

• You can get this for $40 directly from the TI store as a bundle and you can also get it from distributors like Digi-key, Mouser, or Element14 for a similar price. The TI SimpleLink™ MSP432™ microcontroller (MCU LaunchPad development kit is a low cost and modular hardware for developing embedded systems and the MKII Educational BoosterPack™plug-in module is an input and output board that makes prototyping easy with several sensors, human input devices, and a color LCD.

Where can I learn how to use an RTOS?

• TI provides training with SimpleLink Academy. Here, you can learn all about using TI-RTOS for the TI MSP432 MCU LaunchPad development kit. You can also look at text books, online materials and online courses.

Additional Resources:

• Get step-by-step directions for how to teach embedded systems and IoT courses using the TI LaunchPad development kit in the PDF included below
• Download the slides from the 2017 ASEE workshop on Embedded Systems, RTOS and IoT
  
• Find more workshop materials from TI workshops at www.ti.com/asee2017
• Get more university resources at www.university.ti.com 

This post is co-authored by Collin Wells.

Photodiode-based light sensing is a common technique where operational amplifiers (op amps) are used to condition a photodiode sensor for a wide range of applications. An example of this is in smoke detectors, where a photoelectric smoke alarm is used to identify the presence of smoke in the chamber.

A photodiode sensor produces a current proportional to the light level presented to it. Depending on the application, the photodiode is operated in either a photovoltaic or photoconductive mode; each has its own merits, which Bruce Trump discussed in detail in this post from his blog, The Signal.

In a smoke detector system, the photodiode operates in a photoconductive mode, meaning you will typically use a transimpedance amplifier to amplify the photodiode current. In photoconductive mode, the photodiode is held at a zero-volt (Figure 1a) or reverse voltage bias (Figure 1b), preventing it from forward biasing.

Figure 1: Photodiode in photoconductive mode with zero-volt bias (a); or reverse-voltage bias (b)

Equation 1 calculates the direct current (DC) transfer function for the circuits shown in Figure 1 (note that the photodiode current (i_D) is flowing away from the op-amp inverting node): 

The three-step process outlined in John Caldwell’s series on transimpedance amplifiers (see part 3, “What op amp bandwidth do I need?”) determines the minimum required op-amp gain bandwidth for transimpedance configurations. The minimum bandwidth is based on the required transimpedance gain and signal bandwidth, along with the total capacitance presented to the inverting node of the op amp. The diode capacitance often dominates the inverting-node capacitance, but don’t forget to include the effects of the op-amp input capacitance. We summarized the three steps explained in John’s posts here for quick reference.

1. Choose the maximum feedback capacitance (C_F) based on the feedback resistor (R_F) and the signal -3dB bandwidth (f_P) (Equation 2):

2. Calculate the total capacitance (C_IN) at the inverting input of the amplifier. For the circuits shown in Figure 1, this is equal to Equation 3:

where C_J is the diode junction capacitance, C_D is the op-amp differential input capacitance and C_CM2 is the op-amp inverting input common-mode input capacitance.

3. Calculate the minimum required op-amp gain bandwidth product (f_GBW) (Equation 4):

By following these three simple steps, you can avoid many of the stability and performance issues commonly associated with transimpedance amplifier circuits by selecting an amplifier with sufficient bandwidth to perform the required transimpedance gain at the desired signal bandwidth. Op amps with GBWs between 1 - 20MHz are well suited for smoke detector applications because they are able to amplify the low-level signals in the system to sufficient levels while maintaining stability.

Along with meeting bandwidth requirements, the op amp must also meet the system’s DC accuracy requirements. The most important DC specification in many transimpedance applications is the input bias current (i_B) of the op amp. i_B will directly sum or subtract with the input signal current, which can cause large errors depending on the magnitude of i_B compared to the signal current. In smoke detector applications, this DC voltage enables system designers to set the thresholds for the amount of smoke detected before an action is taken in the system.

The example shown in Figure 2 uses a 5MΩ resistor to apply a 5MV/A gain to a 100nA full-scale input current. With the input bias current set to 0A, the full-scale output voltage is 500mV – which is expected based on the transfer function in Equation 1. The circuit on the right in Figure 2 displays the effects of the same circuit with an op amp i_B of 10nA. In this case, the output voltage is 450mV, which shows that the 10nA input bias current caused a 50mV (or 10%) error from the ideal 500mV output signal.

Figure 2: Input bias current effects in transimpedance amplifier circuits

Equation 5 calculates the percentage error of the full-scale range (%FSR) based on the full-scale input current (i_{IN_FS}) and the op amp’s i_B:The TLV6001 device is part of a family of high-performance general-purpose amplifiers for a wide variety of cost-conscious transimpedance applications, such a smoke detectors. This is due to the op amps balance between GBW (1 MHz), low Iq (100 µA), low input bias current (1 pA) and low input capacitance (6pF). Other key features that make this family attractive for system designers in smoke detectors are the RRIO swings and the EMI hardened inputs that help reject interference from extrinsic noise sources.

Table 1 lists different transimpedance gain and bandwidth combinations for the TLV6001 based on Equations 1 through 4. Be sure to keep the total input capacitance below the maximum input capacitance (C_{IN­_MAX}) to avoid stability issues.

Table 1: Quick design calculator for TLV6001 transimpedance applications

Figure 3 shows the simulated step-response results for a 1MV/A gain and 50kHz bandwidth with the maximum 54pF of input capacitance from the photodiode. The output overshoot and ringing are minimal, indicating a stable design.

Figure 3: TLV6001 step-response results; gain = 1MV/A, bandwidth = 50kHz

Smoke detector applications use op amps in the transimpedance configuration to amplify low-level photodiode currents. Designing the transimpedance circuit for smoke detectors can be simplified to a few easy steps. First, follow the three steps from John’s blog posts to select the required op-amp bandwidth. Then sort the remaining results to find a device with an i_B specification that meets the system’s DC requirements. The TLV600x devices highlighted in this blog are a great family of products to design with for cost-sensitive transimpedance applications.

Do you have questions about op amp designs in smoke detectors? Log in and leave a comment below letting us know your experience with transimpedance configurations or any questions you have.

Additional resources

• Find commonly used analog design formulas in the “Analog Engineer’s Pocket Reference” e-book.
• Download The Signal e-book for 32 op amp design lessons that will make you a better engineer.
• See TI’s portfolio of performance op amps for cost-conscious applications.
• Read the other two parts in John Caldwell’s three-part blog series on transimpedance circuits:
  • “Part I – What op amp bandwidth do I need? (Transimpedance Amplifiers).”
  • “Part II – What op amp bandwidth do I need? (Transimpedance Amplifiers).”

Intelligent integrated motor drivers and brushless DC (BLDC) motors can both help electric vehicles and next-generation automobiles become more attractive, viable, and reliable.

Integrated motor drivers combine everything required to drive a motor, such as field-effect transistors (FETs), gate drivers, and state machines, as shown in Figure 1. Integration prevents long routing of wires from the electronic control unit (ECU) to the motor and has additional advantages of smaller printed circuit board (PCB) size and overall system cost.

The advantages that BLDC motors provide in automotive applications include efficiency, compact size, longer motor and battery life, quieter in-cabin experience, and better electromagnetic interference (EMI) performance.

Figure 1: Intelligent integrated BLDC motor driver

In this integrated intelligence blog series, I will describe the different performance requirements for BLDC motors and what makes TI integrated motor drivers “intelligent.” In this first installment, I will elaborate on EMI management in a BLDC system for automotive applications.

BLDC motors are driven at high switching frequency in the range of 10-100kHz. At this high frequency, the combination of high dv/dt and parasitic inductance causes high-frequency ringing on the switching node. This ringing emits high-frequency noise that can interfere with other components in the car.

Adjusting the slew rate of the applied voltage can help reduce interference caused by ringing, as shown in Figures 2 and 3. In a discrete system, adjusting the gate-driver resistor modifies the slew rate of the voltage. You have to change the resistor value manually and select an optimal value based on the test results. The process of manually changing the resistor is tedious and requires multiple iterations of PCBs, which increase both overall size and complexity.

In the case of integrated drivers like the DRV10983-Q1, the gate resistor is not accessible and cannot be changed – and that’s not a bad thing. For example, slew-rate control is integrated in the DRV10983-Q1; you can easily change this slew rate by changing the register value, which speeds up the whole exercise of testing modules for EMI.

Figure 2: Sample EMI measurement with a slew rate of 120V/µs for the DRV10983-Q1 and a BLDC motor

Figure 3: Sample EMI measurement with a slew rate of 35V/µs for the DRV10983-Q1 and a BLDC motor

Another way to improve EMI performance is by changing the pulse-width modulation (PWM) switching frequency. The PWM switching frequency has an effect on ringing.  In case of integrated drivers, this PWM frequency can be changed by configuring register. For example, the DRV10983-Q1 has two frequencies (25kHz and 50kHz) to choose from.

One common technique used to reduce EMI is dithering the main clock frequency. Dithering reduces the amplitude of peak frequency by spreading it across the spectrum.

By using motor drivers with fully integrated features such as slew-rate control, changeable PWM switching frequency and dithering, you can reduce the number of external components for filtering. This saves system costs, board space, and – most importantly – the amount of time it takes to figure out emission sources and the effort of having to redesign boards.

In future blogs, I will discuss startup reliability, initial position detection, anti-voltage surge, resynchronization while a motor is spinning in opposite or same direction, sinusoidal commutation and many other integrated features that make motor drivers intelligent.  

Additional resources

• Download the DRV10983-Q1 data sheet.
• Start spinning those BLDC motors with the DRV10983-Q1 evaluation module.
• Watch a video with advice how to quickly start up the DRV10983-Q1 BLDC motor driver EVM.
• Explore other training videos on a variety of topics that help you design with motor drivers.

As electronics get smaller and smaller, power-supply designers must consider thermal limits when designing their power supplies. A smaller power supply is not useful if it cannot operate at a heavy load inside a specific application environment, which includes the ambient temperature.

One common thermal limit is represented in a derating curve, which you’ll find in most power-module data sheets. The derating curve shows the amount of drawable current or power at various ambient temperatures, while still keeping the power module within its temperature specification (usually below 125°C). Figure 1 shows two such curves from the 2A TPS82140 power-module data sheet.

Figure 1: Derating curves for the 2A TPS82140 power module

As Figure 1 illustrates, derating curves change slightly with changes in input and output voltage, so it is important to look at the appropriate curve for a given design. Generally, derating gets slightly worse as the output voltage increases, because the total output power – and thus the total power losses – are higher. This is counter-balanced by the efficiency, which tends to increase with increasing output voltage, and helps reduce the power loss. Finally, derating curves are based on a specific printed circuit board (PCB), which is usually the power module’s evaluation module (EVM). Unlike the Joint Electron Device Engineering Council (JEDEC) test PCB, the EVM more closely reflects a real-world design.

Pin-to-pin and drop-in compatible with the 3A TPS82130, the 2A TPS82140 and 1A TPS82150 offer much better derating performance, which reduces the power-supply designer’s headaches. Even with a 5V output, the TPS82140 safely gives its full 2A current up to a very balmy 65°C. Figure 2 shows the lower-current TPS82150 supplying its full 1A current up to 95°C. Even here in Texas in the summer, that is downright hot!

Figure 2: Derating curves for the 1A TPS82150 power module

Of course, to get the derating performance shown in the data sheet requires a decent PCB layout. But with just five external passives and a total solution size of about 42mm², a good PCB layout is easy to accomplish.

An easy-to-design, small power module that gets its heat out. Where can you use it in your circuits?

Additional resources

• Check out these other E2E™ Community blog posts:
  • “A smaller step-down power module for communications equipment systems.”
  • “A flexible, easy-to-design MicroSiP power module for portable test and measurement.”
  • “Smaller size now possible in 3A point-of-load converters.”

One solution to the MRI power supply is the LM5140-Q1, an automotive-qualified dual-channel synchronous buck controller. One of the features of the LM5140-Q1 that make it desirable for a MRI application is its ability to be synchronized to an external clock up to 2.6MHz.

The LM5140-Q1 works in many nonautomotive applications because it solves certain specific design challenges. For example, since the device operates at 2.488MHz, you can use it in an MRI power supply.

MRI inductor design steps

The inductance required for an MRI power supply is proportional to the switching frequency, as shown in Equation 1:

where L is inductance in microhenries, V_OUT is the output voltage, ΔI is the inductor ripple current, F_SW is the switching frequency and D is the duty cycle.

Once you have calculated the required inductance, you can use Equation 2 to determine the air-core inductor size:

where L is inductance in microhenries, d is the coil diameter in inches, I is the coil length in inches and n is the number of turns.

Looking at Equations 1 and 2, you can see that a higher switching frequency will result in a lower inductor value. A lower inductance value yields a smaller air-core inductor.

An alternative to the LM5140-Q1 is the LM5141 controller. The LM5141 is the commercial single-channel equivalent of the LM5140-Q1, and has the same features as the LM5140-Q1.

Table 1 lists the typical power-supply requirements for MRI equipment. The highest power rail is 12V at 20.5A, from a 48V (nominal) input. The combination of metal-oxide semiconductor field-effect transistors (MOSFETs) R_DS(ON) and switching losses (which dominate MOSFET losses when operating at 2.488MHz) make thermal management extremely challenging.

The solution is to replace the MOSFETs with gallium nitride (GaN) FETs. GaN FETs provide significant efficiency improvements over MOSFETs because they have nearly zero reverse recovery, lower R_DS(ON) and a lower gate charge (Q_G), reducing the losses to a more manageable level. GaN FETs have critical gate-drive requirements, so the LM5113 GaN FET driver is also necessary.

Table 1: MRI power rails

One of the more challenging design requirements for MRI applications is the need for a negative output voltage at high output currents. This presents another challenge to overcome. In Table 1 are the power requirements for an MRI inverting buck-boost power supply, 48V to -15V (and 48V to -8V), at 15.84A. The inverting buck-boost topology transfer function (Equation 3) requires the LM5140-Q1 to be able to withstand V_IN + V_OUT, 50V_MAX + 15V = 65V.

The LM5140-Q1 is able to operate with an input voltage of 65V (70V absolute maximum), overcoming the danger of overvoltage stresses.

Summary

The most valuable capability of the LM5140-Q1 controller in the context of MRI applications is its ability to be synchronized at 2.488MHz, reducing the size of the air-core inductors and keeps the switch-mode power supply switching frequency outside the sensitive ranges of MRI equipment. This allows for accurate processing of the measured signals in MRI equipment, which is the key to obtaining high quality images.  TI offers a wide variety of products for MRI systems and equipment manufacturers, including op amps, DSPs, multi-channel high- and low-speed data converters, clocking distribution, interface, and power management.