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Let’s say that you’re a power electronics engineer and your boss has asked you to extend the battery life of your product. After optimizing the front-end power path (battery charger) and mid-rail converters (DC/DCs and low-dropout regulators [LDOs]), you believe there’s no room left to squeeze out a few more hours or days of battery life. You’re almost ready to report back that it’s impossible, but after taking a look in your toolbox, you find the solution: load switches.

One of the many ways that you can use a load switch is to reduce the shutdown current of any load. Do you have a Bluetooth® or Wi-Fi® module that consumes over 10µA in deep sleep or hibernation mode? Try adding a load switch like the TPS22916 in Figure 1, which can reduce the shutdown current to just 10nA.

Figure 1: Reduce shutdown current by adding a low-leakage load switch between the supply and load.

In some applications, such as wearables, building automation or medical devices, there can be several sensors and wireless transmitters in a single product that load switches can disable. Figure 2 shows the system block diagram of a typical smartwatch and where load switches would be used in the power path.

Figure 2: Smartwatch system block diagram

Take an example of a smartwatch using a standard 3.7V lithium polymer battery with 65mAh of capacity. In order to last at least five days before charging (a typical work week), you need to leave certain sensors powered (like the step counter), but you can shut down other areas of the board, like the Bluetooth® module. If the Bluetooth® module draws 10µA when disabled, it is contributing to at least 10µA × 24 hours × 5 days = 1.2mAh of the 65mAh budget. In other words, this one module is contributing at least 1.8% (1.2mAh/65mAh) to overall battery-life loss. Plus, this very conservative estimate does not take into account minimum/maximum specs over temperature, nor efficiency loss through the DC/DC converters. If you have several modules leaking current, the situation can multiply very quickly.

How do you combat this? Using the TPS22916 will cut that shutdown leakage current to just 10nA. This means that you can continue using your favorite Bluetooth module and it will have virtually no effect (0.0018%) on battery life when disabled, thanks to your new favorite load switch.

To learn more about load switches and the features they can offer, be sure to check out ti.com/loadswitch.

As technology continues to boom, interconnectivity between devices enables the rapid growth of automated homes. The development of the vacuum robot came about as a result of the convenience of manipulating home equipment through wireless connectivity and remote accessibility.

A vacuum robot eliminates the need for manual floor cleaning by setting schedules to ensure a consistently clean home. In addition to convenience and time savings, these compact vacuum robots – unlike traditional, hefty vacuums – relieve the strain of running a vacuum by easily reaching tough nooks and crannies associated with furniture, walls and corners.

To maximize the floor space cleaned, a vacuum robot must optimize its running time. Thus, a battery-charging solution requires astute consideration in vacuum robot design. Taking an input voltage of approximately 19-20V from a charging dock, most vacuum robots on the market employ rechargeable four-cell lithium-ion (Li-ion) batteries to power their systems. Extending the run time of these batteries and cutting battery costs requires efficient utilization of the battery pack to its fullest capacity. Figure 2 illustrates the vacuum robot system and highlights its battery-management solutions.

Figure 1: Vacuum robot system power diagrams

You can implement the charging circuit in several ways. The discrete solution uses a simple DC/DC converter to charge the battery. The system’s microcontroller (MCU) mimics a CC-CV charging curve by controlling the on/off switching of the MOSFETs. While the discrete solution may be inexpensive, its inaccurate charge voltage, low switching frequency and lack of built-in battery protections contribute to additional costs and poorer performance than a charger integrated circuit (IC).

Alternatively, charger IC solutions offer high charge voltage accuracies, high switching frequencies and enhanced battery protections. While some designers may choose a discrete solution with a lower cost than the charger IC solution, the benefits of a charger IC impressively outweigh the price difference.

Vacuum robots on the market vary in run time, ranging from about 60 to 150 minutes. Maximizing the amount of cleaned surface area relies on the robot’s operational period, making optimal run time a key selling point for consumers; just several extra minutes could make the difference between an entirely clean and partially dirty home.

TI’s charger IC such as the bq24725A or bq24610 offers a high-accuracy charge voltage of ±0.5% compared to a low-cost DC/DC converter charge voltage accuracy of ±5%. Due to the small ±0.5% charge voltage accuracy, this charger IC maximizes battery capacity, which ultimately extends the robot’s run time.

Figure 3 describes the battery voltage versus the depth of discharge (DOD) of a 4.2 Li-ion battery at room temperature. Based on the charge voltage accuracies of the charger IC and several discrete solutions, the data maps several DOD points to battery voltages, which translate to run time. As shown in Figure 3 and its associated data in Table 1, the TI charger IC solution significantly maximizes capacity over discrete charging solutions.

Figure 2: Li-ion DOD versus battery voltage

Table 1: Charge voltage accuracy mapped to battery capacity

Battery capacity ultimately translates to device run time and the cost difference of batteries with a certain run-time goal. For example, take a vacuum robot that uses a TI charger IC solution with ±0.5% charge voltage accuracy that runs for 120 minutes. The same vacuum robot using a DC/DC converter as a discrete solution with a charge voltage accuracy of ±5% would only run for 55 minutes. Therefore, the loss in capacity due to a less accurate charge voltage, as shown in Table 1, significantly diminishes the run time of the robot.

From a monetary standpoint, a battery pack for this application costs about $20. The 56% capacity lost in this example requires you to purchase $11 more in capacity. This extra 65 minutes of run time would allow the robot to clean several additional rooms, and the cost savings due to maximized capacity quantifies the value of using a charger IC solution.

Charger IC solutions yield high switching frequencies and in turn require small, low-cost inductors. For example, TI’s bq24725A, with a switching frequency of 750kHz, typically uses a 4.7µH inductor sized at 28mm². Alternatively, a discrete solution with a switching frequency of only 50kHz requires a larger, 75µH or greater inductor covering about 113mm² of board space. Along with saving solution size, the charger IC’s inductor is roughly two times less expensive than the discrete solution’s inductor, depending on inductor choice.

From a design perspective, a charger IC advantageously offers a complete, sophisticated suite of battery safety features, including input overcurrent, charge overcurrent, battery overvoltage, thermal shutdown, battery shorted to ground, inductor short and field-effect transistor (FET) short protections. On the other hand, a discrete solution would have to use its MCU to implement battery protections, and damage could occur to the battery by the time the MCU detects faults due to its slow response times. Therefore, a charger IC protects the battery in any worst-case scenarios while eliminating the need for you to create your own battery protections.

For additional design flexibility, the TI multicell switching charger portfolio provides options for stand-alone and host-controlled topologies. A stand-alone charger such as the bq24610 controls voltage and current limits with external circuitry elements, facilitating simple implementation. A host-controlled charger such as the bq24725A or bq24773 uses I²C or SMBus to program the limits, saving bill-of-materials cost by employing the calculating power of the system’s already-existing MCU.

A charger IC offers a myriad of advantages over common discrete charging solutions for vacuum robots. Although a discrete solution may be more economical initially, a charger IC solution provides a significantly longer run time, system cost savings and a simpler design implementation than the discrete alternative. Ultimately, the benefits of the complete charger IC solution outweigh the initial cost savings of the discrete charging solution.

Additional resources:

Find resources and reference designs for vacuum robots here.
Explore high efficiency battery charger ICs like bq24725A, bq24773 or bq24610
Don’t forget to consider battery capacity indicators as well. Read this blog for advice on fuel gauges.

If I were to ask you when the Internet of Things (IoT) was born, you’d probably say sometime in the last decade. It’s important to remember, though, that connected devices have been with us for over a half a century. Back in 1981, Hayes Communications introduced a solution to reuse existing cable modem data lines by enabling software that could control a modem’s activity. This solution, later known as ATtention (AT) commands, revolutionized the computer industry and enabled easy and simple man-to-machine and machine-to-machine communication.

AT commands was such a successful solution that it stayed in use throughout the years, surpassing its original use in dial-up modems and continuing to be the de-facto application programming interface (API) standard in modern 2/3/4G cellular modems and other up-to-date communication devices. This means that there is a huge market of connected devices out there that rely on AT commands and need to add IoT connectivity to their systems.

Companies often want to refresh legacy systems by adding new connectivity support options without developing completely new software. Therefore, any activity usually comes with a hefty cost in terms of both time and money. As shown in Figure 1, an alternative solution would be to add Wi-Fi connectivity, by using the AT commands protocol to connect the existing system(s) (software and hardware) to the cloud without being required to do major software and hardware design changes.

Figure 1: Adding IoT capabilities to a legacy system

TI recently released an AT commands software library for the SimpleLink Wi-Fi CC32xx wireless microcontroller (MCU) that fully supports all SimpleLink networking capabilities via AT commands. For customers working with AT commands, this software library will make integration that much easier, directly reducing time to market and the level of investment required.

The CC32xx AT commands library was designed in a modular way and provides much more than the older AT commands interface for networking device capabilities. As shown in Figure 2, it is possible to easily add more capabilities to the AT command library by modifying the AT command application layer.

Figure 2: Basic architecture scheme for AT commands

The SimpleLink Wi-Fi AT commands solution consists of two main modules:

The AT commands core includes the commands parser, command execution and return status handler.
The AT commands application is responsible for managing the interface between the external host processor and the CC32xx device. The default method is to use the universal asynchronous receiver transmitter (UART) interface to communicate with the device. However, due to the modular architecture, it can be just as easy to use the Serial Peripheral Interface (SPI), Secure Digital Input Output (SDIO) or even the networking interface itself to control the device from a remote location. The AT commands software development kit (SDK) provides two built-in options:
- The AT_commands UART provides a UART interface for sending AT commands to the CC32xx network processor.
- Serial_wifi provides the ability to send AT commands to the CC32xx network processor both locally via a UART interface and remotely from a Transmission Control Protocol/Internet Protocol (TCP/IP) interface. This option also enables the connection and control of a remote device using AT commands or can have a remote device control the local device.

As shown in Figure 3, with only four simple serial UART commands, a system can scan a Wi-Fi network, connect to a Wi-Fi router, open a TCP socket and connect to a remote server.

Figure 3: AT commands – UART terminal screenshot

The SimpleLink Wi-Fi AT commands offering is an easy-to-use solution that provides CC32xx platform networking capabilities in a modular and scalable fashion, from simply controlling the CC32xx device from an external MCU located on the same physical platform using the UART interface, to controlling the device from a remote MCU over a TCP cloud connection.

Additional resources

For more information about SimpleLink wireless MCUs and the SimpleLink platform approach, check out the CC3220 SDK.

Let’s start with what might be an all-too-familiar scenario. You find yourself working late and alone; maybe you’re typing out some last emails for the evening or crunching some numbers for an upcoming deadline. Suddenly, the lights in your...(read more)

DLP® technology is gaining popularity in automotive applications such as augmented reality (AR) head-up displays (HUDs) and one of the primary reasons is the bright, vivid colors that it can deliver. To better understand how color plays a role in...(read more)

Many electronic system designers are interested in implementing USB Type-C™ and USB Power Delivery (PD) while getting their products to market as quickly as possible. Most USB Type-C applications require a microcontroller (MCU) because of the need for firmware configuration via the I²C, Serial Peripheral Interface (SPI) and/or Universal Asynchronous Receiver Transmitter (UART) data communication protocols.

But what if you could implement USB Type-C and USB PD without any firmware configuration or an external MCU and get your product to market fast?

Let’s say that you are designing an AC/DC power adaptor with the UCC28740 flyback controller using opto-coupled feedback and the TPS25740B USB Type-C and USB PD source controller, as shown in Figure 1.

Figure 1: AC/DC adapter simplified schematic using USB Type-C and USB PD

The TPS25740B has three control pins (CTL1, CTL2, and CTL3) which adjust the output voltage of the power supply based on the voltage requested by the attached sink. In other words, the CTL pins adjust the resistive feedback network of the optocoupler transmitting to the UCC28740 in order to output the desired voltages on the V_BUS line in real time. This is what enables effortless USB Type-C PD adoption without the need for any firmware implementation.

Table 1 below shows the TPS25740B CTL pin states as a function of the target voltage on V_BUS.

Table 1: TPS25740B CTL states as a function of target voltage on V_BUS

The voltages that are advertised depend on the USB PD source controller and how the device is configured. There are a variety of USB PD source controllers on the market today that can be configured to advertise a range of commonly desired voltages. One family of these devices and their voltage offerings can be seen below in Table 2.

Table 2: TPS25740 and TPS25740X device comparison table

In conclusion, there are a variety of USB Type-C and PD source controllers available today that can be used to reduce time to market. So, consider using a device without the need for firmware or an MCU for your USB Type-C and PD design and get your product to market fast.

Additional Resources

Start your design with these reference designs

Imagine an electric-vehicle charger that gets you on the road twice as fast as chargers used today or a motor drive that takes half the space and offers more efficiency than current applications or a laptop computer power adapter that fits in your pocket.

The future of electronics depends on power-management innovations.

Or consider this: Every simple Internet search query uses enough electricity to burn a 60-watt light bulb for about 17 seconds. Now multiply that by billions of queries happening every day and you end up with billions of kilowatt hours of energy consumption.

The challenge to manage energy more efficiently and squeeze more power into smaller spaces continues unabated. New innovations such as gallium nitride (GaN) promise to significantly improve many aspects of power management, generation and delivery. It’s expected that power electronics will manage about 80 percent of energy by 2030, up from 30 percent in 2005.¹ This amounts to more than 3 billion kilowatt hours of energy savings. That’s enough electricity to power more than 300,000 homes for a year.

Anything that gets its power directly from the grid – from smartphone chargers to data centers – or that deals with high voltages up to hundreds of volts can benefit from technologies such as GaN that will improve the efficiency and size of power-management systems. (Read our new white paper: GaN drives energy efficiency to the next level.)

The search for a perfect switch

The centerpiece of any power-management system is a switch, which turns power on and off. It’s like a light switch on your wall, except millions of times faster and smaller. Efficiency (low losses), reliability, integration and affordability are critical attributes of a semiconductor power switch.

The search for the ideal switch is ongoing. The ideal switch conducts current with little “on” resistance and blocks the current with as little as possible leakage current while blocking significant voltage across its terminals in the off state. A higher switching frequency also means that engineers can design smaller overall power-conversion solutions. Above all, semiconductor switches must be reliable and able to be manufactured cost-effectively.

Silicon power switches have been improving in power efficiency, switching speed and reliability over several decades. These devices have successfully addressed efficiency and switching frequency in low voltage – below 100 volts – or high-voltage tolerance (IGBTs and super-junction devices). However, due to the limitation of silicon, all these features cannot be offered in a single silicon power FET. Wide bandgap power transistors such as GaN and silicon carbide (SiC) promise to offer high power efficiency at high voltages and high switching frequencies above and beyond silicon MOSFET offerings.

What GaN can do for you

An efficient high-frequency switch can reduce the size of power modules by three to 10 times, depending on the application, but requires an optimized driver and controller topology. The totem pole AC/DC converter is a topology, not viable in silicon, that can benefit from GaN’s low on resistance, fast switching and low-output capacitance to offer three times higher power density. Resonant architectures such as zero voltage and zero current switching that mitigate switching losses and improve overall efficiency can also benefit from GaN’s superior switching characteristics.

Many applications require power conversion from relatively high voltage – in the hundreds of volts – to low voltages supplied to circuit components such as processors. Switched-mode power converters with a high input-to-output voltage ratio offer lower efficiencies. These power-management blocks usually involve multiple stages of conversion. Direct conversion from intermediate 54/48 volt bus to the processor core voltage can reduce costs and improve efficiency. GaN, with its unique switching properties, is a strong candidate for direct conversion architectures. Direct conversion is currently being studied for the power management of servers in data center applications.

Also, applications such as laser drivers for LIDAR in autonomous vehicles, wireless charging, and envelope tracking by high-efficiency power amplifiers in 5G base stations can benefit from the efficiency and fast switching of GaN technology.

The reduced conduction loss of GaN power devices, in conjunction with a higher switching frequency, results in much higher power density. But thermal management and parasitics do not scale! Concentrating more power in a smaller volume creates new challenges for heat dissipation and packaging. A smaller die surface area limits the efficiency of traditional packaging techniques. Three-dimensional heat spreading is a promising option for GaN packaging.

Living greener

In order to break the cycle of cost and mass adoption, a new power semiconductor technology needs to address some of the shortcomings of incumbent devices in the most compelling applications. GaN is opening the door to drive power scaling beyond what silicon can offer in high-voltage applications. An inverter for an industrial motor drive or a grid-tied energy storage system can immensely benefit from the higher density offered by GaN devices.

GaN offers other unique, untapped properties that can deliver new values and opportunities for future power management. The bidirectional structure of a GaN device, unlike a typical PN junction MOSFET, can control the current flow with a dual-gate structure. A matrix converter for motor drives can potentially reduce the number of switches by taking advantage of a bidirectional device. Furthermore, GaN devices can operate at higher temperatures than silicon devices, which make it an attractive choice for many hot applications, such as integrated motor drives.

The long-term implications of groundbreaking technologies such as GaN are significant: The lower power loss will mean we won’t need as many new power plants to meet increasing demands for electricity. Higher power density will mean more integration. Battery-powered circuits – such as those in electric vehicles, drones and robots – can run longer and more efficiently. Data centers with their thousands of servers that help us connect with friends and colleagues will operate more efficiently. We’ll be able to live greener lives.

Additional resources:

Check out TI’s portfolio of GaN solutions.
Read our new white paper: GaN drives energy efficiency to the next level
Read our white paper on GaN and SiC power processes: GaN and SiC enable increased power efficiency in power supplies
Read: Optimizing GaN performance with an integrated driver.
Join the TI E2E Community Gallium Nitride Solutions forum to find solutions, get help, share knowledge and solve problems with fellow engineers and TI experts.

1- Power Electronics for Distributed Energy Systems and Transmission and Distribution Applications, ORNL, 2005

Why does your microcontroller (MCU) deserve a nap? Much like humans, MCUs generally consume less energy while sleeping. Designing low-power connected applications is key to extending battery and product life and proliferating the widespread adoption of networked sensors in homes, buildings and cities. The more that your MCU is sleeping, the longer the battery life of your application.

In a wireless system built around single-chip solutions like SimpleLink™ multiband CC1352R or multistandard CC2652R wireless MCUs, the main MCU of the device is tasked with the majority of the computationally intensive work, like running a protocol stack and the majority of the application code. Although the Arm® Cortex-M4F, the main application MCU of the SimpleLink CC26x2R and CC13x2R devices, has excellent power performance while awake (59µA/MHz), the system consumes significantly less power while sleeping (0.8µA). That’s why it is important to let your MCU “nap” between the wakeup periods in which it completes the periodic tasks for which it is optimized. This enables low-power wireless designs to have expanded battery lifetimes in connected applications. But how can you design a useful application if your main MCU is always sleeping?

The SimpleLink sensor controller is a dedicated, 16-bit central processing unit (CPU) core designed to be very low power with respect to active mode, standby mode and startup energy. This core executes code from dedicated ultra-low-leakage (ULL) random access memory (RAM) and includes several low-power analog and digital peripherals (see Figure 1). It can run completely independently while the main Arm Cortex-M4F application processor naps as long as possible before waking up to complete a task. Many low-power wireless sensor applications can benefit from this low-power system architecture.

Figure 1: Multistandard block diagram for CC13X2R and CC26X2R

Take for example a thermostat system in a large commercial building. A central thermostat receives temperature input from remote, battery-powered temperature nodes installed throughout the building to intelligently control the climate in different zones. The CC1352R or CC2652R devices uses the sensor controller’s serial peripheral interface (SPI) to monitor the remote digital temperature sensors, reading temperature data 20 times per second with an average current consumption of 1.4µA (see Figure 2). If the temperature falls outside of a specific range, the main core and radio of the multiband CC1352R will wake up and report the temperature data back to the central thermostat over a Sub-1 GHz network. Using the multiband functionality of the device, the system can also send Bluetooth® low energy alerts to user’s smartphones. With this ultra-low power operation, battery-powered sensor nodes can run for years off of a coin-cell battery, enabling their placement in remote locations throughout a large building.

Figure 2: Sensor controller peripherals

The sensor controller is a key differentiator of SimpleLink multistandard CC26x2R and CC13x2R wireless MCUs, allowing for sleepy wireless application designs. Low-power designs are difficult to achieve, yet they are critical for the widespread adoption of networked sensor applications. With the SimpleLink sensor controller engine, your main MCU can nap and your sleepy wireless application can run for years on a coin-cell battery.

Additional resources

Read this TI TechNote, “Ultra-Low Power Designs With the CC13x2 and CC26x2 Sensor Controller,” for more information.
Buy a LaunchPad™ development kit:
Read the application note, “Getting Started With the CC13xx and CC26xx Sensor Controller.”

These days, it feels like the “portable future” is right around the corner. Devices that used to be cumbersome and bulky have become light and portable. I saw this firsthand in personal electronics: cellphones were once heavy and slow; they’re now slim, fast and have an increasingly longer battery life.

I’ve also witnessed this trend in personal health care. It is now possible to check vital signs without having to go to the doctor, partly because of the increasingly small solution size and low power of devices such as blood glucose monitors in the palm of your hand. Blood glucose monitors are experiencing a growing trend in lower power with longer battery life that enables users to have a responsive body vital measurement device.

Blood glucose monitors are devices that have extremely low power and try to push quiescent current (Iq) to the lowest limits possible because they must be able to measure at least 1,000 tests on the same battery which is typically a lightweight 3V button cell. Reaching a battery life that can handle 1,000 tests has become an issue as blood glucose monitors start to become more connected with Bluetooth® and other wireless connectivity (as shown in Figure 1). This is because the increase in wireless connectivity in turn increases power consumption and lowers battery life and increase the need for multiple coin-cell (220mAh) or even AAA batteries (1,000mAh), which have an increase in size and weight.

Figure 1: Blood glucose monitor system diagram

Unlocking higher accuracy

As system power decreases it can become a challenge to maintain a high accuracy, so it is important to make sure that system accuracy remains high; one way to increase accuracy is to have an external voltage reference. An external voltage reference is often practical because there are several techniques – such as oversampling on an analog-to-digital converter (ADC) – that increase the requirements of the reference voltage well above what a typical internal voltage reference can do. The increased requirements on a voltage reference can be a combination of an improved initial accuracy, a lower temperature coefficient, lower noise or even a different voltage reference level. These requirements are typically difficult to achieve in a lower-power application, but the REF3320 from the REF33xx family of voltage references solves these issues by providing a high accuracy, low temperature coefficient at a low Iq.

A low Iq voltage reference

The REF3320 is one of TI’s low-power precision voltage references. The largest advantage of the REF33xx family is its typical 3.9µA low supply current requirement and its ability to source and sink up to 5mA for an ADC or digital-to-analog converter (DAC) while active. This allows the REF3320 to have a very minor impact on overall system power while the system is sampling, as shown in Figure 2.

Figure 2: Total power consumption overhead example (estimated percentage)

The REF33xx family also offers low-voltage options between 1.25V (the REF3312) and 3.3V (the REF3333) to benefit applications that use coin-cell batteries. These output voltages offer you the flexibility to get the most out of your ADC by selecting an adequate voltage reference to take advantage of the complete full-scale signal. Higher output voltages also give you the flexibility to power ultra-low-power ADCs such as the ADS7042 that rely on using AVDD as the voltage reference.

Low Iq is possible in more ways than one

TI’s large, low-power voltage reference portfolio includes several options aside from the REF33xx family. For certain applications that need the absolute lowest power for regulation, the 1.25V REF1112 has a typical power consumption of 1µA in a small package, making it TI’s lowest-power voltage reference.

But there are more ways to save power. One such way would be to use the enable feature of the REF3425to limit power consumption while the device is active. This is a feature in TI’s REF3425, which is a high-precision voltage reference that can achieve 2.5µA of Iq in shutdown mode. It is also possible to use a load switch to turn of sections of the system to lower Iq which can further bring down the standby current of a blood glucose monitor. Figure 3 shows the power consumption of several popular low-power voltage references.

Figure 3: Typical Iq of several TI voltage references

1.5mm-by-1.5mm UQFN voltage reference

The REF3320 or REF1112 also shine in continuous-sampling low-power monitors such as gas analyzers, personal radiation detectors and smoke detectors. These low-power applications continuously sample every minute (up to several hundred samples) while still maintaining a small battery and form factor. For example, due to the harsh temperature conditions in radiation detectors, the REF3320 has a low temperature drift of 30ppm/°C from -40°C to 125°C that ensures an accurate reference across temperature. In addition, the REF3320 is available in a small-form-factor 1.5mm-by-1.5mm UQFN package, as shown in Figure 4. This small form factor also gives you the flexibility to add passives for additional noise filtering and still be smaller than a typical SOT23-3 package.

Figure 4: The REF33xx family in a UQFN package

Don’t force yourself to choose between low system power and high system accuracy, as it is possible to increase ADC and DAC accuracy by adding a low-power voltage reference like the REF3320 or REF1112. No matter which application you are designing for, TI offers a large portfolio of voltage references that can help you unlock more accuracy in your system.

Additional resources

Read the blog posts on how to reduce power consumption in data-acquisition systems and how to ensure precision in automated processes.
Download the “Tips and tricks for designing with voltage references” e-book.

At the recent Motors & Drive Systems Conference, I discussed electric motor applications in automobiles. The U.K. and France have already set dates to outlaw internal combustion engines (ICE), with China also studying when to ban ICE vehicles. Volvo has already announced that its new cars will have electric drives starting in 2019.

So the writing is on the wall that powerful electric motors, also known as traction motors, will play a significant and increasing role as the engine propelling the vehicle. But electric motors are already dominant in many other automotive applications. Let’s take a motor census of the typical automobile.

Figure 1: Electric motor applications in an automobile

Existing – and increasing – motor populations

Electric starter motors have been part of automobiles since your great-grandparents decided there had to be a better way than a hand-crank to start the car. These are still typically the most powerful electric motors other than traction motors. With the advent of start-stop technology and mild hybrid vehicles, the starter motor is morphing into the starter-generator, and taking on more functions. In some designs, an enhanced starter motor can be used to “creep” forward in stop-and-go traffic, blurring the lines between a starter motor and an electric traction motor.

Windshield wipers are perhaps the most prevalent example of electric motors in existing automobiles. Every car has at least one wiper motor for the front wipers. The popularity of SUVs and hatchbacks with less-streamlined back windows has meant the presence of rear wipers and corresponding motors on a large fraction of cars as well. Another motor pumps washer fluid to the windshields, and in some cars to the headlights, which may have their own small wipers.

Just about every car has blower fans that circulate air from the heating and cooling system; many vehicles have two or more fans in the cabin. High-end vehicles have fans built into the seats for cushion ventilation and heat distribution.

Power seats are fertile ground if you’re looking for electric motors. In economy cars, motors provide convenient front and back adjustment and back cushion tilt. In premium cars, electric motors control options like height adjustment, bottom cushion tilt, lumbar support, headrest adjustment and cushion firmness. Other seat functions that use electric motors include power-seat folding and power stowage of back seats.

Windows used to crank up by hand, but now power windows are common; future generations won’t understand the traditional circular hand motion to ask someone to lower their windows.

Each window is another potential location for an electric motor, including variants such as sunroofs and rear-vent windows in minivans. The drives for these windows can be as simple as a relay, but safety requirements such as detecting an obstacle or pinched object lead to more intelligent drive options, with motion monitoring and limits on drive force.

Locks are another convenience option where manual operation has given way to an electric motor drive. The advantages of electrical control include convenience features such as remote operation, enhanced security and intelligent functions, such as automatic unlock after a collision. Unlike power windows, power door locks must retain the option of manual operation, so this impacts the design of the electric door lock motor and mechanism.

Indicators on the instrument panel, or cluster may evolve to light-emitting diodes (LEDs) or other types of displays, but for now, each dial and gauge uses a small electric motor. Other electric motors in the convenience category include common features like side mirror fold and position adjustment, as well as more exotic applications like convertible roofs, extendable running boards, and glass partitions between the driver and passengers.

Under the hood, electric motors are becoming more common in several places. In most cases, electric motors are replacing belt-driven mechanical components. Examples include radiator fans, fuel pumps, water pumps and compressors. Moving these functions from a belt drive to an electric drive has several advantages. One is that driving electric motors with modern electronics can be much more power-efficient than using belts and pulleys, leading to benefits like higher fuel efficiency, reduced weight and lower emissions. Another advantage is that using electric motors rather than belts allows freedom in mechanical design, as the mounting position of pumps and fans need not be constrained by having to run a serpentine belt to each pulley.

Technology trends

Most electric motors in today’s cars run from the standard 12V automotive system, with a belt-driven alternator to generate voltage and a lead-acid battery for storage. This arrangement has worked fine for decades, but the latest vehicles need more and more current for comfort, entertainment, navigation, driver assistance and safety features.

A dual-voltage 12V and 48V system could move some of the higher-current loads off the 12V battery. The advantages of using a 48V supply are a 4x reduction in current for the same power, and an accompanying reduction in weight in terms of cables and motor windings. Examples of high-current loads that may migrate to a 48V supply include the starter motor, turbocharger, fuel pump, water pump and cooling fans. Implementing a 48V electrical system for these components could result in fuel-consumption savings of around 10 percent.

Brushed DC motors are the traditional solution for driving most electric convenience features in an automotive body. Since the brushes provide the commutation, these motors are simple to drive and are relatively inexpensive. In some applications, brushless DC (BLDC) motors can provide significant benefits in terms of power density, thus reducing weight and providing better fuel economy and lower emissions. Manufacturers are using BLDC motors in windshield wipers, cabin heating, ventilation and air-conditioning (HVAC) blowers and pumps. In these applications the motor tends to run for long periods, as opposed to momentary operation such as in power windows or power seats, where the simplicity and cost-effectiveness of brushed motors still hold an advantage.

Conclusion

So how many electric motors are in your car? You would be hard-pressed to find a late-model car with less than a dozen electric motors, while typical modern cars on American roads might easily have 40 electric motors or more. The increasing popularity of electric vehicles will spur many innovations in automotive electric motors. However, electric motors are already prevalent throughout ICE-propelled vehicles, with more applications in each successive model year bringing more convenience, better intelligence and safer operation while reducing environmental impact. Still – there is always room for more.

Additional resources:

Learn more about electric motors in these blog posts:

TI exhibited the newest automotive exterior lighting solutions at the 13th Auto Lamp Industry Development Technical Forum and Fourth Shanghai International Auto Lamp Exhibition (ALE) on March 28-29 in Shanghai. ALE Shanghai is an annual automotive lamp show at which the top innovators and companies share the most updated technologies around automotive exterior lighting.

TI had three demos at ALE: an automotive rear combination lamp (RCL) taillight demo featuring the TPS92830-Q1 LED controller, a sequential lighting/ambient lighting demo using the TLC6C5712-Q1 LED driver and a next-generation adaptive frontlight system (AFS) light-emitting diode (LED) matrix headlight.

As automotive exterior lighting becomes more sophisticated, most of the new requirements are focused around supporting animations, styling or pixel-brightness control to enhance road safety. Our demos showed the capabilities of TI devices to help customers build on required features in both headlights and RCLs.

As shown in Figure 1, the automotive RCL taillight demo featuring the TPS92830-Q1 LED controller showed different kinds of signal functions that the linear-based TPS92830-Q1 device can support (such as turn, tail and stop), with both analog dimming and pulse-width modulation (PWM) dimming. The demo also highlighted the TPS92830-Q1’s integrated diagnostic and thermal-protection features.

Figure 1: Automotive RCL taillight demo featuring the TPS92830-Q1 LED controller

The sequential lighting/ambient lighting demo using the TLC6C5712-Q1 LED driver exhibited the ability to achieve differentiated styling with red-green-blue (RGB) color mixing (Figure 2). The TLC6C5712 supports diversified automotive applications with independently controlled brightness and color for each LED or lighting bar (through either analog or PWM dimming), and a 12-bit constant-current sink architecture. The device has built-in diagnostic features to improve robustness of the lighting system.

Figure 2: Sequential/ambient lighting using the TLC6C5712-Q1 LED driver

Finally, as shown as Figure 3, the next-generation AFS LED matrix headlight demo enables necessary intelligent headlight functions such as dynamic beam shaping, glare-free high beams, light customization and animation for welcome lights and/or sequential turn lights. It showcases the TI-designed lighting control unit (LCU) as a complete LED driver and lighting management solution for headlights, with the TPS92518HV-Q1 as the constant-current source to drive LEDs and the TPS92662-Q1 as the LED matrix manager to control an individual pixel light’s brightness through shunt-field effect transistor (FET) PWM dimming.

Figure 3: A next-generation AFS LED matrix headlight using the TPS92518HV-Q1 and TPS92662-Q1

TI has many other devices to help build innovative, robust and high-performance solutions in automotive exterior lighting applications. ALE Shanghai is definitely one of the most important events that connects TI and automotive lighting industry gurus. TI is continuing to define, design and make products that are adapting to the challenging standards for automotive exterior lighting needs.

For more information on TI's LED portfolio visit ti.com/LED

The Internet of things (IoT) has implications for Industry 4.0, motivating smart sensor transmitters and similarly capable programmable logic controller (PLC) analog input modules. A previous blog post illustrated how to achieve digital communication by integrating the Highway Addressable Remote Transducer (HART) protocol into a sensor transmitter design. But that’s only half the story. At the other end of the current loop sits the analog input module, which must convert the analog current and HART signals into digital data for processing.

To summarize the previous post, a HART modem couples a frequency-shift keying (FSK) current or voltage signal onto the current loop in a field transmitter. The primary variable (PV) is represented by the DC current value and other data is transferred by the AC FSK waveform. The question is, how do you separate and digitize these signals at the receiver end?

Before designing a HART-enabled analog input module, it is useful to visualize the signal content in the frequency domain. Figure 1 shows the frequency content of a HART-modulated current loop. The analog signal representing the PV is very low frequency, while the HART signal is 1.2kHz-2.2kHz. Figure 1 shows the approximate filtering requirements to separate the signals and avoid interference.

Figure 1: Analog Signal and HART Frequency Bands

Now, you just need a circuit to perform the filtering and convert the signals into the digital domain. Figure 2 shows a HART-enabled 2-wire analog input module schematic. The input module uses the DAC8740H HART modem to demodulate the FSK signal and the ADS8689 to digitize the primary variable.

Figure 2: HART Enabled Analog Input Module Diagram

Let’s look at how the process variable is captured in this circuit. R1 is the load resistor used to measure the current in the loop. The signal is filtered by a resistor-capacitor (RC) low-pass filter with a very low cut-off frequency. This ensures complete attenuation of the AC HART signal, leaving only the process variable. An analog-to-digital converter (ADC) like the ADS8689 then digitizes the shunt voltage and calculates the current in the loop using Equation 1:

(1)

The ADC used to digitize the process variable must have high input impedance in order to avoid altering the effective load resistance, causing errors in the loop-current measurement. The ADS8689 features high input impedance, an integrated analog front end allowing high input voltages and an internal reference.

The denoted HART portion of the circuit consists of the DAC8740H HART modem, with input and output pins AC-coupled to the high side of the load resistor. The DAC8740H’s input (MOD_IN) demodulates the received FSK signal and the output (MOD_OUT) modulates an FSK signal to communicate with the field transmitter. Designers often mistakenly think that a current waveform is used for bidirectional communication in HART, but the field transmitter is regulating the loop current and the input module has no control over the current. Instead, the input module uses a voltage waveform to communicate back to the field transmitter. Since HART is half-duplex, both the input and output can couple to the same node.

You can achieve additional filtering of the HART signal within the DAC8740H to isolate the signal before demodulating. The device contains an internal fourth-order bandpass filter that is enabled by connecting a 680pF capacitor to MOD_INF.

Now you know how to achieve two-way HART communication through the analog input module. For more information about HART, 2-wire sensor transmitters and the devices mentioned in this post, see the links below.

Additional resources

Learn more about TI’s portfolios of precision DACs and ADCs.
Jump-start your design with the highly-accurate, loop-powered, 4mA to 20mA field transmitter with HART® modem reference design.
Read the first blog post in this series, “Back to the basics: what is HART protocol and how does it work?”

The fast-growing consumer electronics market brings opportunities as well as challenges for boost converters. The huge volume drives the market to be very cost-sensitive, so you will need to make trade-offs between solution cost and performance.

The need for true load-disconnect functionality

A boost converter does not have a native mechanism for load disconnection. The rectifier diode or body diode of the synchronous field-effect transistor (FET) passes the battery voltage to the load even when the boost converter is not operating. This leads to continuous battery drainage, even though the leak current may be small.

Many applications require the complete load disconnection and elimination of battery-current leakage when the circuit is not operating. For instance, a boost converter in an electric shaver only needs to operate when it is in use, to boost the battery voltage up for the LED backlight and the motor. Because the electric shaver is off most of the time, disconnecting the loads (the LED backlight and motor in this example) can avoid draining the battery and extend the shaver’s service time between charging or battery replacement.

Obviously, a boost converter with an integrated load-disconnect function is a solution, but the cost of such a device is much higher than a converter without one. This is because a true load disconnect has to be implemented with two high-side power metal-oxide semiconductor field-effect transistors (MOSFETs) in a back-to-back connection, or with a single power MOSFET that can turn off the FET’s body diode. Both implementations can greatly increase the cost of the boost converter integrated circuit (IC). In contrast, a boost converter like the TPS61322xx with an external load disconnect switch offers a cost-effective solution.

True load-disconnect configuration options

You can implement a load-disconnect function by placing an external switch Q1 in the input power path of the boost converter, as shown in Figure 1. A mechanical switch, p-channel FET, p-channel n-channel p-channel (PNP) transistor, n-channel FET or n-channel p-channel n-channel (NPN) transistor can serve as the disconnect switch.

A popular choice is a mechanical switch, which doesn’t require any extra control logic circuit but loses the ability to be controlled by the system microcontroller (MCU). Solid-state semiconductor devices are more controllable and robust, though. P-channel or n-channel FETs are usually preferable to PNP or NPN transistors because the latter two consume continuous base current to drive.

Between the n-channel FET and p-channel PFET, the R_DS(on)of the n-channel FET is less than half the p-channel FET for the same size, and cheaper as well. However, it is challenging to design the printed circuit board (PCB) layout for an n-channel FET because you have to use it to break the ground path. Pay careful attention to the ground routing in these systems. Some applications prohibit the interruption of ground routing, because the broken ground leaves the load circuit energized, which can be a big risk.

Figure 1: Load disconnection by mechanical switch

Another popular solution is to use a p-channel FET on the high side of the power path, which doesn’t interrupt the system ground routing. Figure 2 shows a circuit configuration when the supply voltage of the MCU is higher than the battery voltage, and where the MCU directly controls main switch Q1. If the MCU’s supply voltage is lower than the battery voltage, the general-purpose input/output (GPIO) voltage will not be high enough to turn off Q1 successfully. The solution is to use the configuration shown in Figure 3, where introducing a small n-channel FET or NPN transistor (Q2) helps control Q1.

Figure 2: Load disconnection by p-channel FET, V_MCU > V_Battery

Figure 3: Load disconnection by p-channel FET, V_MCU < V_Battery

If the ground routing is not a concern, an n-channel FET or NPN transistor can fulfill the load-disconnect function, as shown in Figure 4. This approach is simpler than the p-channel FET configuration, and the MCU controls Q1 directly.

Figure 4: Load disconnection by n-channel FET

Switch considerations

It is very important to select a suitable load-disconnect switch. Unlike the MOSFETs of DC/DC converters, the load-disconnect switch is either on or off without frequent switching. Therefore, the gate charge Qg and the parasitic capacitances of the disconnect switch are not a main concern in component selection. Do pay attention to two things, however:

The rated voltage of the switch should be higher than the battery voltage.
Assess the R_DS(on) of Q1 by allowing about 1% total solution efficiency loss under the maximum load. Use Equations 1 and 2 to calculate the power loss of the disconnect switch:

where IIN_RMS is the root-mean-square (RMS) value of the input current and R_DS(on) is the on-resistance of the switch.

As an example, selecting an MOSFET with an R_DS(on)< 25mΩ for 3V-to-5V conversion under a 500mA load causes about 1% loss in overall efficiency. The gate threshold should be less than the minimum battery voltage in order to guarantee operation in the entire range of the battery voltage. Q1 should support in-rush current during startup, during which the battery will charge both the input and output capacitors, and there are not many current-limit factors except for the R_DS(on) of Q1 and the boost converter’s internal synchronous FET. This in-rush current is not only related to the slew rate of Q1’s gate voltage, but also affected by the input and output capacitors.

Test results

I conducted a test with the TPS613222ADBVR, a fixed 5.0V output voltage boost converter. The conditions were V_IN = 1.8V, 2.7V, 3.6V, 4.2V, L = 2.2µH.

As Figures 5, 6 and 7 show, the efficiency differences are very small between the circuit with and without a load-disconnect switch. The worst case is at a heavy load under a low V_IN condition, because if V_IN< 1.8V, Q1 will not have an adequate gate voltage to fully drive the FET. The R_DS(on) will increase and hurt the efficiency.

Figure 5: TPS613222A efficiency without a load-disconnect switch

Figure 6: TPS613222A efficiency with n-channel FET disconnect switch (FDN337N, R_DS(on)= 82mΩ at V_GS = 2.5V)

Figure 7: TPS613222A efficiency with p-channel FET disconnect switch (FDN306P, R_DS(on)= 50mΩ at V_GS = -2.5V)

Conclusion

A device like the TPS613222A provides a cost-effective solution to applications requiring a true load-disconnect function. You can decide to add the appropriate type of switch according to your end-application requirements. True load disconnection is easy to achieve and the total cost can still remain very competitive. For further information, read the blog post, “How to Select a MOSFET – Load Switching.”

(Please visit the site to view this video)

The Huffaire Monoplane featured on the cover of Sports Aviation Magazine in 1969 was a pile of parts and pieces last year, tucked away in storage at an airport hangar classroom in North Texas.

But high school senior Bryan Soltys-Niemann and his classmates in the McKinney Aviation Academy – a four-year vocational aviation program for McKinney Independent School District high schools – dragged it out piece by piece, determined to restore the plane to its former glory.

“We want to be able to say, ‘Look what this used to be,’” Bryan said. “We thought it could be a fun project.”

As students huddled around the plane’s skeletal frame to tinker with the engine, two members of our company’s aviation team looked over their shoulders to lend expert guidance on the mechanics – and to offer real-world advice about engineering their futures in flight.

Paying it forward

Mario Arango, a senior aircraft technician at our company, lit up when he first heard about McKinney Aviation Academy four years ago. The course teaches students everything from the history of aviation to airplane maintenance and operation.

“I was in a similar program when I was in high school, and hearing about this program at the McKinney school district brought back great memories,” he said. “I know how valuable the program was for me. It’s so important to have mentors who can give you real-world advice about how to be a professional in aviation. I’ve had those mentors in the past, and I think the best way to pay it forward is to do the same thing for someone else.”

Mario and Jason Erickson, a pilot for our company, volunteer twice a week to answer students’ questions and help guide the rebuild of the Huffaire Monoplane.

“Mario is the perfect mentor for this project,” said Dan Anderson, our company’s director of aviation. “He understands engines and airframe mechanics, and as a lot of knowledge and expertise that can benefit the kids.”

When Jason learned about the program last fall, he immediately wanted to get involved. He wishes a similar program had existed when he was in high school and before he joined the U.S. Air Force. In class, he sits alongside students during flight simulations to explain what the various graphics mean and to give guidance on what to look for while flying.

“I’m still learning what I can do to help with this program, and I hope to become more involved over time,” he said. “I want to make a difference and help these kids get to where they want to be.”

Hands-on problem solving

Since kindergarten, Bryan knew he wanted to become a pilot.

“As soon as I saw that the McKinney school district had an aviation program, I knew I had to be in it to start off my career a little earlier than everyone else,” Bryan said. “It means a lot that Mario and Jason come over and share their knowledge. Being surrounded by these professionals helps us learn and helps our aviation academy become a little bit stronger.”

Bryce Worley, a high-school junior, also dreams of becoming a pilot and wants to join the military reserves after college. He said the program has opened his eyes to the mechanical side of aviation, with the help of Mario’s expertise.

“A lot of us don’t know the basics,” Bryce said. “Building the plane helps us see what makes it fly – for example, how the stabilator helps gain lift by moving up and down.”

The program has prepared Tom Devine’s 16-year-old son with more than basic aviation knowledge for his plans to join the Air Force. He’s getting hands-on experience he wouldn’t be exposed to otherwise.

“High schools used to have classes like woodworking or metalworking, and that’s kind of fallen by the wayside,” said Tom, who works in our company’s procurement and logistics group. “The kids in this class aren’t just building an airplane. They’re learning to work with their hands, think through problems, follow detailed plans and directions, and work as a team. Those are some of the bigger takeaways they’re going to get from this experience.”

Broadening horizons

The Huffaire Monoplane will never fly again.

But the students plan to make it functional and restore its appearance to the picture-worthy aircraft it was in 1969. The rebuilding process is helping them gain foundational knowledge and understanding of the engineering behind how planes work.

Jason and Mario hope their presence will do more than support the rebuild – their goal is to paint a bigger picture of what a career in aviation can look like.

“When most people think of careers in aviation, they think of commercial airline pilots,” Jason said. “There’s so much more to it than that, and so many more opportunities. I hope that by being here I can broaden their possibilities.”

To accelerate the design process, we have developed something new call Analog Engineer’s Circuits. They are in a "recipe" format that includes step-by-step instructions, basic formulas, schematic diagrams and SPICE simulations. While each circuit is available as a standalone document, we’ve also compiled these “recipes” into two downloadable Analog Engineer’s Circuit Cookbooks. One e-book focuses on ADC circuits, while the other includes operational amplifier (op amp) circuits. By compiling the circuits into e-books, you’ll have a powerful resource at your fingertips – a comprehensive library of sub-circuit ideas that you can easily adapt to meet your specific design needs.

The circuit shown in Figure 1 is a typical high-voltage battery monitor circuit. This example shows an amplifier input that translates a ±20V input signal to a differential ±4.8V for use with an analog-to-digital converter (ADC).

Figure 1: ADC input translation 20V to 5V

Each Analog Engineer’s Circuit contains a step-by-step guide for component selection with equations, as well as example calculations for the given circuit configuration. You can use this procedure and modify it, if needed, to meet your needs. The component selection section for the example circuit covers the gain, amplifier swing limitations and bandwidth limitations of the circuit.

After the component selection section, each Analog Engineer’s Circuit provides TINA-TI™ SPICE simulation results for design verification. In the example above, we provide output swing, AC transfer characteristics, settling time and noise simulations. Figure 2 shows the noise simulation for this design. Some circuits provide measured results as well.

Figure 2: Total noise simulation for figure 1

Each circuit also includes links to support materials that provide additional detailed theoretical background on the different design steps and simulated results. For example, the noise section of the example circuit provides links to a series of TI Precision Labs training videos that explain in detail how to calculate, simulate and measure noise. In addition to being a good technical resource, the support materials keep each circuit short and simple, focusing on the key information.

I encourage you to download our Analog Engineer’s Cookbooks and/or browse a complete list of op amp and ADC circuits.

Let us know if you have questions about, or suggestions for, our Analog Engineer’s Circuits by leaving a comment below or in our Data Converters or Amplifiers support forums in the TI E2E™ Community.

Additional resources

Learn more about TI’s amplifier and data converter portfolios.
Jump-start your design with TI’s reference designs.

The need for true load-disconnect functionality

True load-disconnect configuration options

Figure 1: Load disconnection by mechanical switch

Figure 2: Load disconnection by p-channel FET, V_MCU > V_Battery

Figure 3: Load disconnection by p-channel FET, V_MCU < V_Battery

Figure 4: Load disconnection by n-channel FET

Switch considerations

The rated voltage of the switch should be higher than the battery voltage.
Assess the R_DS(on) of Q1 by allowing about 1% total solution efficiency loss under the maximum load. Use Equations 1 and 2 to calculate the power loss of the disconnect switch:

where IIN_RMS is the root-mean-square (RMS) value of the input current and R_DS(on) is the on-resistance of the switch.

Test results

I conducted a test with the TPS613222ADBVR, a fixed 5.0V output voltage boost converter. The conditions were V_IN = 1.8V, 2.7V, 3.6V, 4.2V, L = 2.2µH.

Figure 5: TPS613222A efficiency without a load-disconnect switch

Figure 6: TPS613222A efficiency with n-channel FET disconnect switch (FDN337N, R_DS(on)= 82mΩ at V_GS = 2.5V)

Figure 7: TPS613222A efficiency with p-channel FET disconnect switch (FDN306P, R_DS(on)= 50mΩ at V_GS = -2.5V)

Conclusion

One of the questions I receive frequently is why there are so many control modes for step-down DC/DC converters and controllers. Whether hysteretic, voltage mode, current mode, constant on time or D-CAP™ mode control (and all of their derivatives), it seems a new one comes out just as we’ve gotten comfortable with the last one.

A few months ago, TI released a new control mode called internally compensated advanced current mode (ACM), which is used in the TPS543B20. This 18V input, 25A DC/DC converter operates at a programmable fixed frequency like current-mode control and does not require loop-compensation components, but employs a feature called asynchronous pulse injection (API) to enable the fast transient behavior of constant on-time control with less output capacitance.

Ultimately, the best control-mode choice depends on the design problems, and that is the answer to the question that I posed in the title. TI is active in the development of leading-edge control circuits to help engineers address tomorrow’s design challenges. Figure 1 shows 12 different control modes used by nonisolated DC/DC converters and controllers from Texas Instruments.

Figure 1: Control modes for nonisolated step-down controllers and converters

In 2011, I was nominated to monitor control modes of nonisolated DC/DC converters and controllers for TI, and it has become an interesting hobby! There were more than 10 different control modes, including control modes from National Semiconductor. Six years later, there are now several new ones, and I maintain a short training presentation and quick reference guide to help differentiate one control mode from another. Each control mode can take hours to effectively present, so the quick reference guide provides useful links to more technical documentation on the TI website. To find products with a particular control mode more easily, our parametric search for step-down converters features a control-mode parameter.

My other role is to act as the control-mode institutional memory, and my journey started in 1999 when TI released a hysteretic controller for the server market, powering the main motherboard processor. The hysteretic controller improved the transient response time with less output capacitance than current- or voltage-mode control, saving board space and cost. But some designers were apprehensive about using a hysteretic nonlinear controller for the first time in their design. A few years later, derivatives of hysteretic control such as constant on-time and adaptive on-time became available. Designers did not have to spend time taking Bode plots or compensating feedback loops with external compensation components as they did with current mode and voltage mode. We were gaining traction. Designers that used simpler internally compensated current- or voltage-mode converters were satisfied with the constant on-time control modes, actually. The limitations of the inductor and output capacitor versions were undesirable, however, as they provided no means to adjust the loop. When higher-value ceramic capacitors like 47µF and 100µF became more available at lower costs, new derivatives of nonlinear control modes came out to support the low equivalent series resistance (ESR) of ceramic capacitors and provide the tighter reference-voltage accuracy that processors require.

On the other hand, many designers still preferred a linear, predictable, fixed-frequency control, since their applications used high-speed clocks, data converters and noise-sensitive analog circuitry such as those found in the industrial and communications markets. Over the years, several derivatives to these linear control modes were released to allow low conversion ratios with a high input voltage and varying line input voltages.

Again, the best control-mode choice depends on the design problems at hand. Check out our quick reference guide and watch our short training presentation and let us know about your favorite control mode.

Universal Serial Bus (USB) is an industry standard that defines cable, connector and communication protocols for facilitating the connection, communication and power-supply exchange between computers and devices. The use of USB in cars initially arose because consumers wanted an easy way to charge consumer electronics, but more recent applications of USB handle data transmission around the car. There are three types of connectors for USB ports: A, B and C. The type of port connector depends on whether the system port is downstream or whether the USB cable is carrying data. Type A is used solely for hub to hub (Figure 1).

Figure 1: USB Type A connector

TI has developed a 21W dual automotive USB charger solution (Figure 2), which is typical for modern automobiles today. It contains all of the necessary front-end protection against issues like double-battery, reverse-battery and load-dump conditions (compliant with International Organization for Standardization [ISO] 7630 Pulse 1, 2a, 3a/b) and support for deep cold-crank conditions. The 21W dual automotive USB design supports 2 x 2.1A for each of the USB ports and can charge two tablets or smartphones.

The Comité International Spécial des Perturbations Radioélectriques (CISPR) 25 Class 5 Rated Automotive USB Charger Reference Design with Full Front End Protection supports strict automotive electromagnetic interference (EMI) standards like CISPR 25 Class 5, and has a differential EMI filter to support conductive EMI suppression. This is particularly important, as any noise at the system level generated by this design could affect other electronic automotive systems. For example, poor filtering could cause a buzzing noise on the audio system.

Figure 2: 21w Dual Port USB charging solution

The reference design connects directly to the automobile’s 12 V lead-acid battery and includes transient protection circuitry, an EMI filter and reverse-battery protection followed by the LMS3655-Q1 (for voltage regulation), the TPS2561-Q1 USB charging controller and the TPS2561-Q1 for dual-channel current limiting and power switching.

We designed the LMS3655-Q1 to support very low EMI switching noise by controlling the switch slew rate on the regulator, optimizing printed circuit board (PCB) layout and component placement, using jittering techniques like spread spectrum, reducing parasitic capacitors and inductance in the package. By controlling the fundamental EMI generation and limiting the switch node over ringing, it is possible to limit wideband noise generation. See Figure 3.

Figure 3: Switch-node LMS3655-Q1; note the absence of ringing

By controlling the noise generation, it is possible to more easily handle EMI at the system level. By not generating the problem, you don’t need to then solve the problem later with PCB layout or by adding more components.

To verify the results, we tested our design in an EMI chamber under the documented conditions laid out within the CISPR 25 standard; see Figure 4. This design addresses one of the fundamental issues seen in other designs, which was noise leaking from the switch node to the edge connector. By limiting the EMI of the switch node in the design, there was no need for an additional metal shield for suppression. Having a design without such a component is not only more cost-effective but is also smaller and thinner, with the potential ability to run cooler.

Figure 4: EMI setup with PCB, without shielding

Running a design cooler not only has power-saving benefits; it also limits the additional noise associated with high temperatures. Heat can limit the operating life of any system, and automotive systems have to survive many more years than your average consumer electronic device.

The LMS3655-Q1 operates at a sub-AM band to maximize efficiency, but TI also designed it at the integrated circuit (IC) level to operate at the highest level of integration, thermal performance and EMC through a combination of internal field-effect transistor (FET) selection, dead-time reduction and other techniques to minimize parasitic resistance. See Figure 5.

Figure 5: Efficiency curves for the LMS3655-Q1

USB designs are becoming smaller and integrating more features into a defined area. There’s less PCB copper for thermal dissipation for each of the individual component, and the plastic material used to contain the USB designs can add to self-heating. The LMS3655-Q1 offers high efficiency to minimize self-heating inside the design, also will operate at higher temperatures when there are challenging thermal limitations. Offering high levels of integration to minimize size, and features to minimize noise and solution cost. It can be used to overcome these challenges and assist the engineer to achieve a competitive solution. Get more information on TI’s products for automotive applications.

Read additional blogs on the topic of USB:

Conventional transient voltage suppression (TVS) diodes have been essential building blocks in system designs for decades. They protect against transient surge events, and many designers rely on TVS diodes to provide an adequate level of surge immunity to protect sensitive electronics at the heart of their systems.

Many end applications, from industrial sensors and programmable logic controller (PLC) input/output (I/O) modules to appliances and medical equipment, must comply with international standards for surge immunity such as International Electrotechnical Commission (IEC) 61000-4-5.

Although conventional TVS diodes have been the safe option to ensure robust and reliable surge immunity, they are far from perfect. In fact, most designs exposed to harsh environments that require surge immunity are likely to be over-designed due to inefficient clamping and the inherent performance variability of conventional TVS protection diodes across temperature. You must settle for solutions with larger package sizes, excessive line capacitance and higher static leakage current. And you must carefully select downstream components to ensure the residual voltage and current from a surge event doesn’t damage downstream components and lead to system failure.

Because TVS diode technology hasn’t changed in decades, many designers simply reuse their TVS protection schemes from generation to generation. But, it’s time to pull out your schematics and look for a better surge protection alternative. TI’s flat clamps are changing the game, providing improved surge protection with more flexibility so that you can optimize for cost, size and performance.

Figure 1: Clamping comparison between flat-clamp technology and traditional TVS diodes

Would a robust surge protection solution – with as much as a 90% smaller package footprint (as shown in Figure 2), 50% lower static leakage current and significantly lower capacitance – get you to evaluate your TVS protection scheme? What if the clamping voltage was more precise, clamping at a consistent, flat voltage across the duration of the surge event and saving you the cost of high-voltage-tolerant downstream components?

Figure 2: Flat-clamp packages can have a 90% area reduction on the printed circuit board

It’s easy to ignore TVS diodes when redesigning your system and reuse a conventional TVS in your next-generation design. However, robust surge protection can’t be ignored, and it’s time to reconsider that portion of the schematic. There is a better alternative with improved clamping performance that also saves board space and downstream system costs.

Additional resources

Read TI’s whitepaper describing the new flat-clamp surge-protection technology.
Start designing now with the TVS3300 evaluation module.

Over the course of this series of blogs, I have covered a number of different FET applications, from motor control to power supply. Alas, I have arrived at the final topic – selecting a MOSFET for hot swap.

When a power supply is abruptly disconnected from its load, large current swings across the circuit’s parasitic inductive elements can generate dramatic voltage spikes that can be detrimental to the electrical components on the circuit. Similar to a battery protection application, here the MOSFET acts to insulate input power supply from the rest of the circuitry. In this case however, the FET is not meant to immediately sever the connection between input and output, but rather limit the severity of those destructive current surges. This is accomplished via a controller regulating the gate-to-source bias on a MOSFET placed in between an input power supply (V_IN) and the output voltage (V_OUT) forcing the MOSFET to operate in a saturation mode, and thus impeding the amount of current that can pass through (see Figure 1).

Figure 1: Simplified hot swap circuit

Before anything else, your first consideration for this FET should be selecting an appropriate breakdown voltage, which is generally 1.5 to 2x the maximum input voltage. For instance, 12V systems tend to implement 25V or 30V FETs, and 48V systems tend to implement 100V or in some cases 150V FETS. The next consideration should be the MOSFET’s safe operating area (SOA) – a curve provided in the datasheet that is especially useful for indicating how susceptible a MOSFET is turn thermal runaway during short power surges not unlike those it must absorb in hot-swap applications. I wrote a post last year about how we at TI go about measuring and then generating the SOA of a MOSFET as it appears on the device’s datasheet. If you haven’t read that, you may want to consider giving it a skim, because for this application, the SOA is the most crucial criteria for making an appropriate selection.

The critical question for you, the designer, to ask is what is the maximum current surge that the FET might see (or be expected to limit to the output) and how long will this surge last. Once you know this, it is relatively simple to look up the corresponding current and voltage differential on the SOA figure in the device datasheet.

For instance, if your design has an input of 48V and you want to limit the current to the output to no more than 2A for 8ms, you could refer to the CSD19532KTT, CSD19535KTT and CSD19536KTT SOAs’ 10ms curves (Figure 2) and deduce that the latter two devices might work, whereas the CSD19532KTT would be insufficient. But since the CSD19535KTT is good enough with some margin, the performance of the more costly CSD19536KTT may be overkill for this application.

Figure 2: The SOA of three different 100V D2PAK MOSFETs

In the example above, I assumed an ambient temperature of 25˚C, the same condition at which the SOA was measured at on the datasheet. But if the end application could be exposed to a much hotter environment, you must derate the SOA in proportion to how close the higher ambient temperature is to the FET’s maximum junction temperature. Let’s say for instance that the max ambient temperature of the end system is 70˚C; you would derate the curves of the SOA using Equation 1:

In this case, the CSD19535KTT’s 10ms, 48V capability would decrease from ~2.5A to ~1.8A. You would then deduce that particular FET would probably no longer be capable enough for this application and instead, select the CSD19536KTT.

It’s worth noting this method of derating assumes that the MOSFET will fail at exactly the max junction temperature, which is generally not the case. Say the failure points measured in the SOA testing actually occur at 200˚C or some other arbitrary higher value; the calculated derating will be closer to unity. That is to say, this derating methodology errs on the conservative side.

The SOA will also dictate the type of MOSFET package you select. D2PAK packages can house large silicon die, so they are very popular for higher-power applications. Smaller 5mm-by-6mm and 3.3mm-by-3.3mm quad flat no-lead (QFN) packages are preferable for lower-power applications. For current surges less than 5 – 10A, the FET is most often integrated with the controller.

A few final caveats:

While I was speaking specifically to hot-swap applications here, you could apply the same SOA selection process to any situation where the FET operates in the saturation region. You could even use the same method for selecting a FET for OR-ing applications, power over Ethernet (PoE), or even slow switching applications like motor control, where there is going to be substantial high V_DS and I_DS overlap during MOSFET turn off.
Hot swap is an application that tends to use surface-mount FETs as opposed to through-hole FETs (like TO-220s or I-PAK packages). The reason is that the heating that takes place for short pulse durations and thermal runaway events are very localized. In other words, the capacitive thermal impedance elements from the silicon junction to case prevent heat from dissipating into a board or heat sink fast enough to cool the junction. Junction-to-case thermal impedance (R_θJC), a function of die size, is important, but junction-to-ambient thermal impedance (R_θJA), a function of package, board and system thermal environment, is much less so. For that same reason, it is pretty rare to see heat sinks used for these applications.
Designers often assume that the lowest-resistance MOSFET in a catalog will have the most capable SOA. There is some logic to this – lower resistance within the same silicon generation is usually indicative of a larger silicon die inside the package, which does yield greater SOA capability and lower junction-to-case thermal impedance. However, as silicon generations improve in resistance per unit area (RSP), they tend to increase in cell density. The denser the cell structure inside the silicon die, the more susceptible the die tends to be to thermal runaway. That is why older-generation FETs with much higher resistance sometimes also have much better SOA performance. The takeaway is that it always pays to investigate and compare the SOA.
I would be remiss if I did not remind you (as I did in my blog post last year) not all datasheet SOAs are created equal, and you should not take all vendors’ SOAs at face value. Although TI certainly stands by the SOAs in its datasheets, it’s always best to have real data when possible.

TI has a number of hot swap controllers that you can learn more about here. For reference, tables 1-3 at the end of this post highlight some devices we commonly recommend for hot swap, that provide some easy look up values for SOA capability.

That brings me to the conclusion of this MOSFET selection blog series. Thank you for reading, and I hope you learned something in the process. If you have any questions about this or any other post, feel free to post a comment below, or ask a question in TI’s E2E™ Community forum here.

Table 1: MOSFETs for 12V hot swap

MOSFET	V_DS(V)	Package	Typ R_DS(ON)(mΩ)	SOA Current Rating (A) @ 14V V_DS
MOSFET	V_DS(V)	Package	@ 10V V_GS	1ms	10ms
CSD17575Q3	30	SON3.3x3.3	1.9	4.5	2
CSD17573Q5B	30	SON5x6	0.84	8	4.5
CSD17576Q5B	30	SON5x6	1.7	8	4
CSD16556Q5B	25	SON5x6	0.9	25	6
CSD17559Q5	30	SON5x6	0.95	30	14
CSD17556Q5B	30	SON5x6	1.2	35	12
CSD16401Q5	25	SON5x6	1.3	100	15
CSD16415Q5	25	SON5x6	0.99	100	15

Table 2: MOSFETs for 24V hot swap

MOSFET	V_DS(V)	Package	Typ R_DS(ON)(mΩ)	SOA Current Rating (A) @ 30V V_DS
MOSFET	V_DS(V)	Package	@ 10V V_GS	0.1ms	1ms	10ms	100ms
CSD18531Q5A	60	SON5x6	3.5	28	9	3.8	0.9
CSD19502Q5B	80	SON5x6	3.4	30	9	3.2	1
CSD18532NQ5B	60	SON5x6	2.7	100	8.6	3	1.9
CSD18540Q5B	60	SON5x6	1.8	105	13	4.9	2.2
CSD19535KTT	100	D2PAK	2.8	130	18	5.1	3
CSD19505KTT	80	D2PAK	2.6	200	18.5	5.3	3.4
CSD18535KTT	60	D2PAK	1.6	220	21	6.1	4.1
CSD18536KTT	60	D2PAK	1.3	220	31	9.5	5
CSD19506KTT	80	D2PAK	2.0	310	29	10	5.3
CSD19536KTT	100	D2PAK	2.0	400	34	10.5	5.4

Table 3: MOSFETs for 48V hot swap

MOSFET	V_DS(V)	Package	Typ R_DS(ON)(mΩ)	SOA Current Rating (A) @ 60V V_DS
MOSFET	V_DS(V)	Package	@ 10V V_GS	0.1ms	1ms	10ms	100ms
CSD19531Q5A	100	SON5x6	5.3	10	2.7	0.85	0.27
CSD19532Q5B	100	SON5x6	4.0	9.5	3	1	0.33
CSD19532KTT	100	D2PAK	4.6	41	3.3	0.8	0.5
CSD19535KTT	100	D2PAK	2.8	46	6.1	1.9	1
CSD19536KTT	100	D2PAK	2.0	120	11	3.7	1.9

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